Title of Invention

VIRTUAL-FLUX DECOUPLING HYSTERESIS CONTROLLER FOR MAINS CONNECTED INVERTER SYSTEMS

Abstract

(EN) The invention relates to a method for operating a converter circuit. To this end, the converter circuit comprises a converter unit (1) with a multitude of controllable power semiconductor switches and an energy store circuit (2) formed by two series-connected capacitors. The controllable power semiconductor switches are controlled by means of a control signal (S) formed from a hysteresis signal vector (x) and the hysteresis signal vector (x) is formed from a differential phase connection current vector (?ifi,i) by means of a hysteresis controller (6), and the differential phase connection current vector (?ifi,i) is formed by subtracting a phase connection current vector (ifi,i) from a reference phase connection current vector (ifi,i,ref), and the reference phase connection current vector (ifi,i,ref) is formed from a differential output value (Pdiff), a differential reactive power value (Qdiff) and of a phase flow vector (?g,aß). In order to keep the switching frequency of the controllable power semiconductor switches as constant as possible, a current correction value (io) is additionally subtracted for forming the differential phase connection current vector (?ifi,i), the current correction value (io) being formed by integrating a phase connection voltage mean value (Uinv,A), and the phase connection voltage mean value (Uinv,A) is formed by determining the arithmetic mean of the phase connection voltages (Uinv,iM) with a reference point of the connection point (M) of the capacitors of the energy store circuit (2). The invention also relates to a device with which the method can be carried out in a particularly simple manner. (DE) Es wird ein Verfahren zum Betrieb einer Umrichterschaltung angegeben, wobei die Umrichterschaltung eine Umrichtereinheit (1) mit einer Vielzahl an ansteuerbaren Leistungshalbleiterschaltern und einen durch zwei in Serie geschaltete Kondensatoren gebildeten Energiespeicherkreis (2) aufweist, bei dem die ansteuerbaren Leistungshalbleiterschalter mittels eines aus einem Hysteresesignalvektor (x) gebildeten Ansteuersignals (S) angesteuert werden und der Hysteresesignalvektor (x) aus einem Differenzphasenanschlussstromvektor (&Dgr;ifi,i) mittels eines Hysteresereglers (6) gebildet wird und der Differenzphasenanschlussstromvektor (&Dgr;ifi,i) aus der Subtraktion eines Phasenanschlussstromvektors (ifi,i) von einem Referenzphasenanschlussstromvektor (ifi,i,ref) gebildet wird, wobei der Referenzphasenanschlussstromvektor (ifi,i,ref) aus einem Differenzwirkleistungswert (Pdiff), einem Differenzblindleistungswert (Qdiff) und einem Phasenflussvektor (ψg,&agr;&bgr;) gebildet wird. Zur weitestgehenden Konstanthaltung der Schaltfrequenz der ansteuerbaren Leistungshalbleiterschalter wird zur Bildung des Differenzphasenanschlussstromvektor (&Dgr;ifi,i) zusätzlich ein Stromkorrekturwert (io) subtrahiert, wobei der Stromkorrekturwert (io) durch Integration eines Phasenanschlussspannungsmittelwertes (Uinv,A) gebildet wird und der Phasenanschlussspannungsmittelwert (Uinv,A) durch Ermittlung des arithmetischen Mittelwertes der Phasenanschlussspannungen (Uinv,iM) mit Bezugspunkt des Verbindungspunktes (M) der Kondensatoren des Energiespeicherkreis (2) gebildet wird. Ferner wird eine Vorrichtung angegeben, mit welcher das Verfahren in besonders einfacher Weise durchgeführt werden kann.

Full Text

Method for operating a converter circuit, and device for carrying out the method DESCRIPTION Technical Field The invention relates to the field of power electronics, and in particular to a method for operating a converter circuit, and to an device for carrying out the method, as claimed in the precharacterizing clause of the independent claims. Prior Art Conventional converter circuits have a converter unit with a multiplicity of controllable power semiconductor switches, which are switched in a known manner in order to switch at least two switching voltage levels. Furthermore, by way of example, an LCL-filter can be connected to each phase connection of the converter unit. Figure 1 shows one embodiment of a device for carrying out a method for operating a converter circuit according to the prior art. The converter circuit in Fig.1 has a converter unit 1: The converter unit 1 shown in Figure 1 is connected to an energy storage circuit 2 which is formed in the conventional manner by two series-connected capacitors. In order to operate the converter circuit, a device is provided which has a control device 15 for producing a hysteresis signal vector x, which device 15 is connected to the controllable power semiconductor switches in the converter unit 1 via a control circuit 3 for forming a control signal S from the hysteresis signal vector X. The povi/er semiconductor switches are therefore controlled by means of a control signal S. The hysteresis signal vector x is formed by means of a hysteresis regulator 6 from a difference-phase connection current vector Aifj.i. The difference-phase connection current vector Aifij is in turn formed from subtraction of a phase connection current vector ifij from a reference phase connection current vector is,i,ret, with the reference phase connection current vector ifi,i,ref being formed from a reference power value Pdiir, a reference wattless-component value Qdiff and a phase flux vector \(;g,ap. The method for operating a converter circuit described above is subject to the problem that the switching frequency of the power semiconductor switches varies to a very major extent as a result of the formation of the reference phase connection cun-ent ifj.i ref from the reference power value Pref, from the reference wattless component value Qref and from the phase flux vector v|/g,ap. A switching frequency which is variable to such a major extent results in a significant increase in the harmonics in the phase connection currents ifg.i and in the phase connection voltages Uinvj on the converter circuit. In this context, Figure 4 shows a corresponding time profile of a phase connection current ifg,i which is subject to major harmonics, for one phase. By way of example, when the converter circuit is connected to an electrical interconnected power supply system, such high harmonic components are undesirable or are unacceptable. For example, when the converter circuit is connected to an electrical load, harmonics such as these can lead to damage or even destruction, and are therefore very highly undesirable. Description of the Invention One object of the invention is therefore to specify a method for operating a converter circuit, by means of which the switching frequency of controllable power semiconductor switches in a converter unit in the converter circuit can be kept virtually constant. A further object of the invention is to specify an device by means of which the method can be carried out in a particularly simple manner. These objects are achieved by the features of claim 1 and claim 9, respectively. Advantageous developments of the invention are specified in the dependent claims. The converter circuit has a converter unit with a multiplicity of controllable power semiconductor switches, and an energy storage circuit formed by two series-connected capacitors. In the method according to the invention for operating the converter circuit, the controllable power semiconductor switches are now controlled by means of a control signal which is formed from a hysteresis signal vector, with the hysteresis signal vector being formed from a difference-phase connection current vector by means of a hysteresis regulator, and with the difference-phase connection current vector being formed from the subtraction of a phase connection cun-ent vector from a reference phase connection cun-ent vector. The reference phase connection current vector is furthermore formed from a reference power value, a reference wattless component value and a phase flux vector. According to the invention, a current correction value is additionally subtracted in order to form the difference-phase connection current vector, with the current correction value being formed by integration of a phase connection voltage mean value, with the phase connection voltage mean value being formed by determining the arithmetic mean value of the phase connection voltages with the reference point of the connection point of the capacitors in the energy storage circuit. The current con-ection value formed in this way allows the switching frequency of the controllable power semiconductor switches in the converter unit to be advantageously kept virtually constant. The very largely constant switching frequency also advantageously allows the harmonics in the phase connection currents and in the phase connection voltages of the converter unit to be kept low. The device according to the invention for carrying out the method for operating the converter circuit has a control device which is used to produce a hysteresis signal vector and is connected to the controllable power semiconductor switches via a control circuit for fonning the control signal, with the control device having a hysteresis regulator in order to form the hysteresis signal vector from the difference-phase connection current vector, a first adder in order to form the difference-phase connection current vector from the subtraction of the phase connection current vector from the reference phase correction current vector, and a first calculation unit in order to form the reference phase connection current vector from the reference power value, the reference wattless component value and the phase flux vector. Furthermore, the current correction value is also supplied to the first adder in order to form the difference-phase connection current vector, in order to form the difference-phase connection current vector from the subtraction of the phase connection current vector and the current con-ection value from the reference phase connection current vector. Furthermore, the control device has an integrator in order to form the current correction value by integration of a phase connection voltage mean value, and an averager in order to form the phase connection voltage mean value by determining the arithmetic mean value of the phase connection voltages with the reference point of the connection point of the capacitors in the energy storage circuit. The device according to the invention for carrying out the method for operating a converter circuit can then be produced very easily and cost-effectively, since the circuit complexity can be kept extremely low and, furthermore, only a small number of components are required to construct it. The method according to the invention can therefore be earned out particularly easily by means of this device. These and further objects, advantages and features of the present invention will become obvious from the following detailed description of preferred embodiments of the invention, and in conjunction with the drawing. Brief Description of the Drawings In the figures: Fig. 1 shows one embodiment of a device for carrying out a method for operating a converter circuit according to the prior art. Fig. 2 shows a first embodiment of a device according to the inventioh for carrying out the method according to the invention for operating the converter circuit, Fig. 3 shows a second embodiment of a device according to the invention for carrying out the method according to the invention for operating the converter circuit. Fig. 4 shows a time profile of a phase connection current for one phase in a method for operating the converter circuit according to the prior art, Fig. 5 shows a time profile of the phase connection current for one phase using the method according to the invention, and Fig. 6 shows one embodiment of a fifth calculation unit. The reference symbols used in the drawing and their meanings are listed in a summarized form in the list of reference symbols. In principle, identical parts are provided with the same reference symbols in the figures. The described embodiments represent examples of the subject matter of the invention, and have no restrictive effect. Approaches to Implementation of the Invention Figure 2 shows a first embodiment of a device according to the invention for carrying out the method according to the invention for operating a converter circuit. As shown in Figure 2, the converter circuit has a converter unit 1 with a multiplicity of controllable power semiconductor switches, and an energy storage circuit 2 formed by two series-connected capacitors. By way of example, the converter unit 1 in Figure 2 is shown as having three phases. It should be mentioned that the converter unit 1 may in general be designed as any converter unit 1 for switching two or more switching voltage levels (multi-level converter circuit) with respect to the voltage of the energy storage circuit 2 connected to the converter unit .1. In the method according to the invention for operating the converter circuit, the controllable power semiconductor switches in the converter unit 1 are now controlled by means of a control signal S formed from a hysteresis signal vector x. A look-up table is used in the normal manner to form the control signal, in which hysteresis signal vectors x are associated in a fixed fonn with con-esponding control signals S, or a modulator which is based on pulse- width modulation. It should be noted that all the vectors with the index i have vector components corresponding to the total of i phases, that is to say if i=3 phases, the corresponding vectors also have i=3 vector components. The hysteresis signal vector x is also formed from a difference-phase connection current vector Aifjj by means of a hysteresis regulator 6, and the difference-phase connection current vector Aif,,i is in turn formed from the subtraction of a phase connection cun-ent vector if,,i from a reference phase connection current vector ifi,i,ref, with the reference phase connection current vector ifi,i,ref being formed from a difference power value Pdiff, a difference wattless component Qdiff and a phase flux vector vj/g op. The vector components of the phase connection current vector ifij are typically measured by means of current sensors at the appropriate phase connections of the converter unit 1. According to the invention, a current correction value io is additionally subtracted in order to forni the difference-phase connection current vector Ainj, and the current con-ection value io is formed by integration of a phase connection voltage mean value Uinv/i, with the phase connection voltage mean value umv^ being formed by determining the arithmetic mean valufe of the phase connection voltages Uinvjw with the reference point of the connection point M of the capacitors in the energy storage circuit 2. The current correction value io means that the switching frequency of the controllable power semiconductor switches in the converter unit 1 can advantageously be kept virtually constant. The very largely constant switching frequency in tum allows the hannonics in the phase connection cun-ents ifi,i and in the phase connection voltages Umv.i of the converter unit 1 to advantageously be kept low. In this context, Figure 5 shows a time profile of the phase connection current ifi,i for one phase when using the method according to the invention, showing that a considerable reduction can be achieved in the harmonics in the profile in comparison to the profile shown in Figure 4 for a conventional method. The phase flux vector Vg ap is preferably formed from the phase connection current vector ifjj, from the control signal S and from an instantaneous DC voltage value UDC in the energy storage circuit 2. This will be described in detail in the foreign text. It should be noted that all the vectors with the index ap as vector components have an a-component of the space vector transformation of the corresponding variable, and a p-component of the space vector transformation of the corresponding variables. The space vector transformation is in general defined as follows: y = ya + jyp  and ya=-j|(yi-|y2+y3) ^|(y2-y3) where y is a complex variable, Va is the a-component of the space vector transformation of the variable y and yp is the p-component of the space vector transformation of the variable y, and yi, y2, ys are vector components of the vector y associated with the complex variable y. All of the already mentioned space vector transformations of variables and those which will be mentioned In the following text are produced using the formulae quoted above, in which case these can be produced separately in a calculation unit provided specifically for this purpose, or else in the calculation unit in which the corresponding a-components and p- components are required for calculation of another variable. The phase flux vj/g is obtained in a general form, using complex notation as: Uinv,a=-^UDc-fi( TjJ, = -dt-Lg -ifi,! where and •f2(S) Uinv.p -1 and Lg is a power supply system inductance and fi(S), f2(S) are predetermineable switching functions of the control signal S. The formulae mentioned above can therefore be used to form the phase flux vector v|/g,ap, that is to say in particular its components vj/g a, H/g.p, in a very simple form.  According to the embodiment shown in Figure 2, the difference power value Pdiff corresponds to a predetermineable reference power value Pref, and the difference wattless component value Qdiff corresponds to a predetermineable reference wattless component value Qref. In addition to the hysteresis regulator 6 which has been mentioned in order to form the hysteresis signal vector x from the difference-phase connection current vector Aifij, the control device 15 in the device shown in Figure 2 also has a first adder 16 in order to form the difference-phase connection current vector Ainj from the subtraction of the phase connection current vector ifij from the reference phase connection current vector ifi,i,ref, and a first calculation unit 5 in order to form the reference phase connection current vector ifij.ref from the difference power value Pdiff, the difference wattless component value Qdiff and a phase flux vector \|/g.ap. According to the invention, in order to form the difference-phase connection current vector Ainj the first adder 16 is additionally supplied with the current correction value io in order to form the difference-phase connection current vector Aif,,! from the subtraction of the phase connection current vector if,,i and the current correction value io from the reference phase connection current vector ifi i ref. Furthermore, the control device 15 shown in Figure 2 has, according to the invention, an integrator 8 in order to form the current correction value io by integration of the phase connection voltage mean value Uinv,A> and an averager 7 in order to form the phase connection voltage mean value Uinv,A by determining the arithmetic mean value of the phase connection voltages Uinvjw with the reference point of the connection point M of the capacitors in the energy storage circuit 2. The device according to the invention for carrying out the method for operating the converter circuit can therefore be produced very easily and cost-effectively, since the circuit complexity can be kept extremely low and, furthermore, only a small number of components are required to construct it. The method according to the invention can accordingly be carried out particularly easily by means of this device. As shown in Figure 2, the control device 15 has a second calculation unit 4 in order to form the phase flux vector \{;g,ccp from the phase connection current vector ifi i, from the control signal S and from the instantaneous DC voltage value UDC from the energy storage circuit 2. Figure 3 shows a second embodiment of a device according to the invention for carrying out the method according to the invention for operating the converter circuit. In this figure, an LCL-filter Lf.i, Cfi, Lfgj is connected to each phase connection of the converter unit 1. Once again, the index i represents the number i of phases. Accordingly, each LCL-filter has a first filter inductance Lf, a second filter inductance Lfg and a filter capacitance Cf, with the first filter inductance Lf being connected to the associated phase connection of the converter unit 1, to the second filter inductance Lfg and to the filter capacitance Cf. The filter capacitances Cf of thfe individual LCL-filters are also connected to one another. According to the method, in the embodiment shov/n in Figure 3, the difference power value Pdiff is formed from subtraction of a damping power value Pd from the sum of a reference power value Pref and at least one compensation harmonic power value Pn with respect to the fundamental frequency of the filter output current vector ifgj of the LCL-filter, with the damping power value Pd being formed from a sum, weighted with a variable damping factor kd, of a multiplication of aaa-component of the space vector transfonnation of filter capacitance currents icfa of the LCL-filter by an a-component of the space vector transformation of the components of the phase connection cun-ent vector ifjj, and multiplication of a p-component of the space vector transformation of filter capacitance currents ictp of the LCL-filter by a p-component of the space vector transfomnation isp of the components of the phase connection current vector ifij. The difference wattless component value Qdiff is, furthermore, formed from subtraction of the sum of a reference wattless component value Qref and at least one compensation harmonic wattless component value Qh with respect to the fundamental frequency of the filter output current vector ifgj of the LCL-filter, with the damping wattless component value Qd being formed from a difference, weighted with the variable damping factor kd, of a multiplication of the p-component of the space vector transformation of the filter capacitance currents ictp of the LCL-filter by the a-component of the space vector transformation of the components of the phase connection current vector ifj,,, and multiplication of the a-component of the space vector transfonnation of the filter capacitance currents icfa of the LCL-filter by the p-component of the space vector transformation kp of the components of the phase connection cun-ent vector ifij. As shown in Figure 3, the filter capacitance currents are measured by means of current sensors at the corresponding filter capacitances Cfj and are vector components of the filter capacitance current vector icf.i as illustrated in Figure 3. The damping power value Pd is formed using the following formula: Pd -'^d '('Cfa "'fia +'cfp "'fip) The reference power value Pref as shown in Figure 3 is freely variable, and is the nominal value of the power which is intended to be produced at the output of the LCL-filter. The damping wattless component value Qd is formed using the following formula: Qd = l^d ■ ('cfp ■ 'fia ~ 'ati ■ 'fip ) The reference wattless component value Qref as shown in Figure 3 is freely variable, and is the nominal value of the wattless component which is intended to be produced at the output of the LCL-filter. It should be noted that the formation of the damping power value Pd and of the damping wattless component value Qd can be avoided by just calculating a damping current vector from the a-component of the space vector transformation of filter capacitance cun-ents icfa of the LCL-filter and from the p-component of the space vector transformation of filter capacitance currents icfp of the LCL-filter by suitable filtering, which damping current vector is then included directly in the formation of the reference phase connection current vector ifi,i,ref and therefore in the formation of the difference-phase connection current vector Aifi,,. This is associated with a saving of computation time, since there is advantageously no need to calculate the damping power value P^ and the damping wattless component value Qd. The damping power value Pd and the damping wattless component value Qd advantageously make it possible to actively dampen distortion, that is to say undesirable oscillations, in the filter output currents ifgj and filter output voltages, so that this distortion is greatly reduced and, best of all, is very largely suppressed. A further advantage is that there is no need to connect a discrete, space-consuming complex and therefore expensive damping resistance at the respective phase connection, in order to allow the undesirable distortion to be effectively damped. The addition or connection of at least one compensation harmonic power value Ph in order to form the difference power value Pdifr and of at least one compensation harmonic wattless component value Qh to form the difference wattless component value Qdiff advantageously result in active reduction of harmonics, and therefore, overall, in a further improvement in the reduction of harmonics. As shown in Figure 3, the control device 15 has a second adder 13 in order to form the difference power value Pdiff from the subtraction of the damping power value Pd from the sum of a reference power value Pref and at least one compensation harmonic power value Pn with respect to the fundamental frequency of the filter output current vector i,g,i of the LCL-filter, with the control device 15 having a third calculation unit 9 in order to form the damping power value Pd from the sum, weighted by a variable damping factor kd, of a multiplication of the a-component of the space vector transformation of filter capacitance currents icfa of the LCL- filter by the a-component of the space vector transformation ka of the components of the phase connection current vector ifi j, and multiplication of the p-component of the space vector transformation of filter capacitance currents ictp of the LCL-filter by the p-component of the space vector transformation ifip of the components of the phase connection current vector ifj.i. Furthermore, the control device 15 has a third adder 14 in order to form the difference wattless component value Qdiff from the subtraction of the sum of a reference wattless component value Qref and at least one compensation harmonic wattless component value Qh with respect to the fundamental frequency of the filter output current vector ifgj of the LCL- filter, with the third calculation unit 9 additionally being used to form the damping wattless component value Qd from a difference, weighted with the variable damping factor kd, of a multiplication of the p-component of the space vector transformation of filter capacitance currents icfp of the LCL-filter by the a-component of the space vector transformation ifia of the components of the phase connection current vectors inj and multiplication of the a-component of the space vector transformation of filter capacitance currents icfa of the LCL- filter by the p-component of the space vector transformation inp of the components of the phase connection current vector ifi,j. It is also feasible for the damping power value Pd and the damping wattless component value Qd to be formed just from the a-component of the space vector transformation of filter capacitance currents iaa of the LCL-filter with an a-component of the space vector transformation ifia of the components of the phase connection current vector ifi.i and multiplication of a p-component of the space vector transformation by filter capacitance currents icrp of the LCL-filter. According to Figure 3, a compensation wattless component value Qcomp is also added in order to form the difference wattless component value Qoiff, with the compensation wattless component value Qcomp being formed by low-pass filtering of an estimated filter capacitance wattless component value Qcf. This advantageously avoids undesirable wattless components of the LCL-filter, in particular of the filter capacitances Cf,i of the LCL-filter, being produced at the output of the LCL-filter, so that it is possible to ensure that only a wattless component value corresponding to the selected reference wattless component value Qref is produced at the output of the LCL-filter. As shown in Figure 3, the compensation wattless component Qcomp is also supplied to the third adder in order to form the difference wattless component value Qdiff, with the compensation wattless component value Qcomp being formed by low-pass filtering of an estimated filter capacitance wattless component value Qcf by means of a low- pass filter 12. The estimated filter capacitance wattless component value Qcf is furthermore formed from the a-compor>ent of the space vector transformation of the filter capacitance currents icfa, from the p-component of the space vector transformation of the filter capacitance currents ictp, from an estimated filter capacitance fiux vector \(;cf,(xp and from the fundamental frequency angle ©t with respect to the fundamental frequency of the filter output current vector ifgj, as is illustrated in particular by the following formula. Qcf =®-(Vcfa-'cfa+VCf|5->Cfp) In order to form the estimated filter capacitance wattless component Qct, the control device 15 has, as shown in Figure 1, a fourth calculation unit 10 by means of which the estimated filter capacitance wattless component value Qcf is calculated using the above formula. The estimated filter capacitance flux vector ycup is formed, as shown in Figure 3, from the instantaneous DC voltage value Udc in the energy storage circuit 2, from the control signal S, from the a-component of the space vector transformation isa of the components of the phase connection current vector if,,i and from the p-component of the space vector transformation iflp of the components of the phase connection cun^ent vector ifij. The second calculation unit 4 is therefore additionally used to form the estimated filter capacitance flux vector \^;cf,ap from the instantaneous DC voltage value UDC in the energy storage circuit 2, from the control signal S, from the a-component of the space vector transformation of the components of the phase connection current vector ifij and from the p-component of the space vector transformation inp  of the components of the phase connection current vector if,,i. The a-component of the space vector transformation \|icfa of the filter capacitance flux vector H'cf.ap is therefore formed using the following formula: UJcfa = |Uinv,adt-Lfifia In a corresponding manner, the p-component of the space vector transformation \j/cfp of the filter capacitance flux vector vi/cf.ap is formed using the following formula: IMcfp = Juinv.pdt-Lf -ifip As shown in Figure 3, the already mentioned compensation harmonic power value Ph and the compensation harmonic wattless component value Qh are each formed from the a-component of the space vector transformation of the filter output currents ifga, from the P-component of the space vector transfomiation of the filter output currents ifgp, from an a-component of the space vector transformation of the filter output fluxes \\iia, from a p-component of the space vector transformation of the filter output fluxes ^Lp and from the fundamental frequency angle cot with respect to the fundamental frequency of the filter output current vector ifg.i. fga The a-component of the space vector transformation of filter output fluxes v|;La is formed from an a-component of the space vector transformation of estimated filter capacitance fluxes ycfa and from the a-component of the space vector transformation of filter output currents ifga, in particular as illustrated by the following formula. VFla =¥cfa-Lfg-i Furthermore, the p-component of the space vector transformation of filter output fluxes \|/Lp is formed from a p-component of the space vector transformation of estimated filter capacitance fluxes yctp and from the p-component of the space vector transformation of filter  output currents ifgp, in particular as illustrated by the following formula. M^LP =VCfp-Lfg-ifgp The a-component of the space vector transformation of filter output fluxes \|;La and the (3-component of the space vector transformation of filter output fluxes vlp can be calculated, for example, in the second calculation unit 4 or can also be calculated in the fifth calculation unit 11, although this is not illustrated for the sake of clarity, in Figure 3. The control device 15 has a fifth calculation unit 11 in order to form the compensation harmonic power value Pn and the compensation harmonic wattless component value Qh in each case from the a-component of the space vector transformation of filter output currents ifga, from the p-component of the space vector transformation of filter output currents ifgp, from the a-component of the space vector transformation of filter output fluxes \\iLa, from the p-component of the space vector transformation of filter output fluxes \|/LP and from the fundamental frequency angle cot with respect to the fundamental frequency of the filter output current vector ifg.i. The filter output current vector ifgj is calculated very easily from the phase connection current vector ifij and from the filter capacitance current vector icf.i, as shown in Figure 3. The fundamental frequency angle cot is provided to the calculation units 9, 10 and 11 from the second calculation unit 4, as shown in Figure 3 from a phase locked loop (or PLL for short) of the second calculation unit 4. Figure 6 shows one embodiment of the fifth calculation unit 11. As shown in Figure 6, the a-component of the space vector transformation of filter output currents itga and the p-component of the space vector transformation of filter output currents Ifgp are first of all formed by the space vector transformation of filter output current vector ifgj that is supplied, in the fifth calculation unit 11. After this, the a-component of the space vector transformation of filter output currents ifg„ and the p-component of the space vector transformation of filter output currents ifgp are Park- Clarke-transformed, or low-pass filtered and are emitted as a d-component and the q- component of the Park-Clarke-Transformation of at least one desired selected harmonic of the filter output currents ihd, ihq with respect to the fundamental frequency of the filter output currents ifgi, ifg2, ifgs- The index h represents the h-th harmonic of this variable and the variables mentioned in the following text, where h=1, 2, 3, ... . The Park-Clarke-Transformation is in general defined as: where a is a complex variable, ad is the d-component of the Park-Clarke-Transformation of the variables a and aq is the q-oomponent of the Park-Clarke-Transformation of the variable a . Advantageously, in the Park-Clarke-Transformation, not only is the fundamental frequency of the complex variable a transformed, but also all of the harmonics that occur in the complex variable a . As shown in Figure 6, the d-component and the q-component of the Park-Clarke-Transformation of the desired selected h-th harmonic of the filter output currents >hd, ihq are each regulated at an associated predetermineable reference value i*hd, i*hqi preferably on the basis of a Proportional-lntegral-Characteristic, and are then inverse-Park- Clarke-transformed, as a result of which an a-component of the space vector transformation of the h-th harmonic of the reference filter output currents i*ha and a p-component of the space vector transformation of the h-th harmonic of the reference filter output currents i*hp are formed. Finally, the compensation harmonic power value Ph and the compensation harmonic wattless component value Qn are each calculated from the a-component of the space vector transformation of the h-th harmonic of reference filter output currents i*ha, the p- component of the space vector transformation of the h-th harmonic of reference filter output currents i*hp, the a-component of the space vector transformation of filter output fluxes vi^La and from the p-component of the space vector transformation of filter output fluxes v|;LP, in particular as illustrated by the following formulae: Ph =®-(VLa-'*hp-VLp-i*ha) Qh =0)-(VLa-'*ha+M^Lp-'*hp) All the steps in the method according to the invention may be implemented as software, which can then be loaded and run, for example, on a computer system, in particular with a digital signal processor. The digital delay times that occur in a system such as this, in particular for the calculations, may, for example, be taken into account in a general form by addition of an additional term to the fundamental frequency cot in the Park-Clarke- Transformation. Furthermore, the device according to the invention as described in detail above may also be implemented in a computer system, in particular in a digital signal processor. Overall, it has been possible to show that the device according to the invention, in particular as illustrated in Figure 2 and Figure 3, for carrying out the method according to the invention for operating the converter circuit can be implemented in a very simple form and cost effectively, since the circuit complexity is extremely low and, furthermore, only a small number of components are required to construct it. The method according to the invention can therefore be carried out particularly simply by means of this device. List of Reference Symbols 1 Converter unit 2 Energy storage circuit 3 Control circuit 4 Second calculation unit 5 First calculation unit 6 Hysteresis regulator 7 Averager 8 Integrator 9 Third calculation unit 10 Fourth calculation unit 11 Fifth calculation unit 12 Low-pass filter 13 Second adder 14 Third adder 15 Control device 16 First adder PATENT CLAIMS 1. A method for operating a converter circuit, with the converter circuit having a converter unit (1) with a multiplicity of controllable power semiconductor switches and having an energy storage circuit (2) formed by two series-connected capacitors, in which the controllable power semiconductor switches are controlled by means of a control signal (S) formed from a hysteresis signal vector (x), and the hysteresis signal vector (x) is formed from a difference-phase connection current vector (Aifjj) by means of a hysteresis regulator (6), and the difference-phase connection current vector (Ainj) is formed from the subtraction of a phase connection current vector (isj) from a reference phase connection current vector (if,,i,ref), with the reference phase connection current vector (ifi,i,ref) being formed from a difference power value (Pdiff), a difference wattless- component value (Qdiff) and a phase flux vector (v|/g,ap), characterized in that a current correction value (io) is additionally subtracted in order to form the difference-phase connection current vector (Aifij) in that the cun^ent correction value (io) is formed by integration of a phase connection voltage mean value (Uinv,A)> and in that the phase connection voltage mean value (umvA) is formed by determining the arithmetic mean value of the phase connection voltages (Uinvju/i) with the reference point of the connection point (M) of the capacitors in the energy storage circuit (2). 2. The method as claimed in claim 1, characterized in that the phase flux vector (vi/g,,^) is formed from the phase connection current vector (inj), from the control signal (S) and from an instantaneous DC voltage value (UDC) of the energy storage circuit (2). 3. The method as claimed in one of claims 1 or 2, characterized in that an LCL-filter (Lfj, Cf„ Lfg.i) is connected to each phase connection of the converter unit (1), in that the difference power value (Pdiff) is formed from subtraction of a damping power value (Pd) from the sum of a reference power value (Pref) and at least one compensation harmonic power valve (Ph) with respect to the fundamental frequency of the filter output current vector (ifg,,) of the LCL-filter, with the damping power value (Pd) being formed from a sum, weighted by a variable damping factor (kd) of a multiplication of an a- component of the space vector transformation of filter capacitance cun-ents (icfa) of the LCL-filter by an a-component of the space vector transformation (if,J of the components of the phase connection current vector (ifij), and multiplication of a p-component of the space vector transformation of filter capacitance currents (icfp) of the LCL-filter by a p- component of the space vector transformation (ifip) of the components of the phase connection current vector (ifij), in that the difference wattless component value (Qdiff) is formed from subtraction of the sum of a reference wattless component value (QreO and at least one compensation harmonic wattless component value (Qh) with respect to the fundamental frequency of the filter output current vector (ifgj) of the LCL-filter, with the damping wattless component value (Qd) being formed from a difference, weighted by the variable damping factor (kd), of a multiplication of the p-component of the space vector transformation of filter capacitance currents (ictp) of the LCL-filter by the a-component of the space vector transformation (isa) of the components of the phase connection current vector (ifij) and multiplication of the a-component of the space vector transformation of filter capacitance currents (icta) of the LCL-filter by the p-component of the space vector transformation (ifip) of the components of the phase connection cun-ent vector (ifij). 4. The method as claimed in claim 3, characterized in that, in order to form the difference wattless component value (Qdiff) a compensation wattless component value (Qcomp) is also added, with the compensation wattless component value (Qcomp) being formed by low-pass filtering of an estimated filter capacitance wattless component value (Qcf). 5. The method as claimed in claim 4, characterized in that the estimated filter capacitance wattless component value (Qcf) is formed from the a-component of the space vector transformation of the filter capacitance currents (icfa), from the p-component of the space vector transformation of the filter capacitance currents (icfp), from an estimated filter capacitance flux vector (\|/cf,ap) and from the fundamental frequency angle (cot) with respect to the fundamental frequency of the filter output current vector (ifgj). 6. The method as claimed in claim 5, characterized in that the estimated filter capacitance flux vector (xj/CF.op) is formed from an instantaneous DC voltage value (UDC) of the energy storage circuit (2), from the control signal (S), from the a-component of the space vector transformation (!«„) of the components of the phase connection current vector (ifij) and from the p-component of the space vector transformation (if,p) of the components of the phase connection current vector (ifij). 7. The method as claimed in one of claims 3 to 6, characterized in that the compensation harmonic power value (Ph) and the compensation hamrionic wattless component value (Qh) are each formed from the a-component of the space vector transfonnation of the filter output currents (ifga), from the p-component of the space vector transformation of the filter output currents (ifgp), from an a-component of the space vector transfonnation of the filter output fluxes (vj/La), from a p-component of the space vector transformation of the filter output fluxes (vup) and from the fundamental frequency angle (cot) with respect to the fundamental frequency of the filter output cun-ent vector (ifgj). 8. The method as claimed in claim 1, characterized in that the difference power value (Pdiff) corresponds to a predetermineable reference power value (Pref), and in that the difference wattless component value (Qdiff) corresponds to a predetermineable reference wattless component value (0^0- 9. A device for carrying out a method for operating a converter circuit, with the converter circuit having a converter unit (1) with a multiplicity of controllable power semiconductor switches and having an energy storage circuit (2) formed by two series-connected capacitors, having a control device (15) which is used to produce a hysteresis signal vector (x) and is connected via a control circuit (3) for forming a control signal (S) to the controllable power semiconductor switches, with the control device (15) having a hysteresis regulator (6) for forming the hysteresis signal vector (x) from a difference-phase connection current vector (Aifi.j), a first adder (16) for forming the difference-phase connection current vector (Aisj) from the subtraction of a phase connection current vector (ifjj) from a reference phase connection current vector (ifi,i,ref) and a first calculation unit (5) for forming the reference phase connection current vector (is,i,ret) from a difference power value (Pdiff), a difference wattless component value (Qdiff) and a phase flux vector (yg.ap), characterized in that a current correction value (io) is additionally supplied to the first adder (16) in order to form the difference-phase connection current vector (Aifij), in order to form the difference-phase connection current vector (Aifjj) from the subtraction of the phase connection current vector (inj) and the current correction value (io) from the reference phase connection current vector (ifi,i,ref), in that the control device (15) comprises an integrator (8) for forming the current correction value (io) by integration of a phase connection voltage mean value (Ujnv,A)> and an averager (7) for forming the phase connection voltage mean value (Uinv/^) by determining the arithmetic mean value of the phase connection voltages (Ujnv.iM) with the reference point of the connection point (M) of the capacitors in the energy storage circuit (2). 10. The device as claimed in claim 9, characterized in that the control device (15) has a second calculation unit (4) for forming the phase flux vector (vj/g,a(}) from the phase connection current vector (if,,i), from the control signal (S) and from an instantaneous DC voltage value (UDC) of the energy storage circuit (2). 11. The device as claimed in claim 9 or 10, characterized in that an LCL-filter (Lfj, Cn, Lfgj) is connected to each phase connection of the converter unit (1), in that the control device (15) has a second adder (13) in order to form the difference power value (Pdiff) from the subtraction of a damping power value (Pd) from the sum of a reference power value (Pref) and at least one compensation harmonic power value (Ph) with respect to the fundamental frequency of the filter output current vector (ifgj) of the LCL-filter, with the control device (15) having a third calculation unit (9) in order to form the damping power value (Pd) from a sum which is weighted with a variable damping factor (kd) of a multiplication of an a-component of the space vector transformation of the filter capacitance currents (icfa) of the LCL-filter by an a-component of the space vector transformation (isa) of the components of the phase connection current vector (ifi.i) and multiplication of a (3-component of the space vector transformation of filter capacitance currents (ictp) of the LCL-filter by a p-component of the space vector transformation (inp) of the components of the phase connection current vector (if,,i), in that the control device (15) has a third adder (14) in order to form the difference wattless component value (Qdiff) from the subtraction of the sum of a reference wattless component value (Qref) and at least one compensation harmonic wattless component value (Qh) with respect to the fundamental frequency of the filter output current vector (ifg,i) of the LCL-filter, with the third calculation unit (9) additionally being used to form the damping wattless component value (Qd) from a difference, which is weighted with the variable damping factor (k^) between a multiplication of the p-component of ^e space vector transformation of filter capacitance currents (icrp) of the LCL-filter by the a-component of the space vector transformation (isa) of the components of the phase connection current vector (if,,i), and multiplication of the a-component of the space vector transformation of filter capacitance currents (icfa) of the LCL-filter by the |3-component of the space vector transformation (if,p) of the components of the phase connection current vector (ifi,i). 12. The device as claimed in claim 11, characterized in that, in order to form the difference wattless component (Qdiir), the third area is additionally supplied with a compensation wattless component (Qcomp), with the compensation wattless component value (Qcomp) being formed by low-pass filtering of an estimated filter capacitance wattless component value (Qcf) by means of a low-pass filter (12). 13. The device as claimed in claim 12, characterized in that the control device (15) has a fourth calculation unit (10) in order to form the estimated filter capacitance wattless component value (Qcf) from the a-component of the space vector transformation of the filter capacitance currents (icfa), from the p-component of the space vector transformation of the filter capacitance currents (ictp), from an estimated filter capacitance flux vector (n/cf.ap) and from the fundamental frequency angle (cot) with respect to the fundamental frequency of the filter output current vector (ifgj). 14. The device as claimed in claim 12, characterized in that, the second calculation unit (4) is additionally used to form the estimated filter capacitance flux vector (vj/cf.ap) from an instantaneous DC voltage value (UDC) of the energy storage circuit (2), from the control signal (S), from the a-component of the space vector transformation (If,J of the components of the phase connection current vector (ifij) and from the p-component of the space vector transformation (ifjp) of the components of the phase connection current vector (ifij). 15. The device as claimed in one of claims 11 to 14, characterized in that the control device (15) has a fifth calculation unit (11) in order to form the compensation harmonic power value (Ph) and the compensation harmonic wattless component value (Qn), in each case from the a-component of the space vector transformation of the filter output cun-ents (ifga). from the p-component of the space vector transformation of the filter output currents (ifgp), from an a-component of the space vector transformation of filter output fluxes (v]/La). from a p-component of the space vector transfomiation of filter output fluxes (v|/Lp) and from the fundamental frequency angle (cot) with respect to the fundamental frequency of the filter output current vector (ifg i). 16. The device as claimed in claim 9, characterized in that the difference power value (PdHr) corresponds to a predetermineable reference power value (Pref), and in that the difference wattless component value (Qoiff) con-esponds to a predetermineable reference wattless component value (Qret)-

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2495-CHENP-2008 AMENDED CLAIMS 23-04-2014.pdf

2495-CHENP-2008 CLAIMS.pdf

2495-CHENP-2008 CORRESPONDENCE OTHERS 18-11-2014.pdf

2495-CHENP-2008 CORRESPONDENCE OTHERS.pdf

2495-CHENP-2008 DESCRIPTION (COMPLETE).pdf

2495-CHENP-2008 DRAWINGS.pdf

2495-CHENP-2008 EXAMINATION REPORT REPLY RECEIVED 23-04-2014,.pdf

2495-CHENP-2008 FORM-1.pdf

2495-CHENP-2008 FORM-18.pdf

2495-CHENP-2008 FORM-3.pdf

2495-CHENP-2008 FORM-5.pdf

2495-CHENP-2008 PCT.pdf

2495-CHENP-2008 POWER OF ATTORNEY.pdf

2495-CHENP-2008 ABSTRACT.pdf

2495-CHENP-2008 AMENDED CLAIMS 10-10-2014.pdf

2495-CHENP-2008 AMENDED CLAIMS 16-12-2014.pdf

2495-CHENP-2008 AMENDED PAGES OF SPECIFICATION 16-12-2014.pdf

2495-CHENP-2008 CORRESPONDENCE OTHERS 16-12-2014.pdf

2495-CHENP-2008 CORRESPONDENCE OTHERS 18-12-2013.pdf

2495-CHENP-2008 EXAMINATION REPORT REPLY RECIEVED 10-10-2014.pdf

2495-CHENP-2008 FORM-1 16-12-2014.pdf

2495-CHENP-2008 CORRESPONDENCE OTHERS 03-12-2014.pdf

4754-CHENP-2008 CORRESPONDENCE OTHERS 28-05-2014.pdf

4754-CHENP-2008 FORM-3 28-05-2014.pdf