Title of Invention

METHOD AND APPARATUS FOR IMPROVING CHANNEL ESTIMATION IN PRESENCE OF SHORT SPREADING CODES

Abstract

A method and apparatus for estimating a communication channel impulse response h(t) is disclosed. The method comprises the steps of generating com(t) = co(t + mNTcc) for m = 0,1,Lambda , M by correlating a received signal r(t) with a spreading sequence Si of length N (208), wherein the received signal r(t) comprises a chip sequence cj applied to a communication channel characterizable by an impulse response h(t)(204), and wherein the chip sequence ci is generated from a data sequence di spread by the spreading sequence Si (202); generating an estimated communication channel impulse response hM (t) as a combination of com(t) and dm for m = 0,1, Lambda ,M (210); and filtering the first estimated communication channel impulse response hM (t) to generate the estimated communication channel impulse response h(t) with a filter f selected at least in part according to the spreading sequence Si (212).

Full Text

METHOD AND APPARATUS FOR IMPROVING CHANNEL ESTIMATE IN PRESENCE OF SHORT SPREADING CODES BACKGROUND OF THE INVENTION 1. Field of the Invention [0001] The present invention relates to systems and methods for communicating information, and in particular to a system and method for estimating the impulse response of a communication channel using short synchronization codes. 2. Description of the Related Art [0002] In packet-based communication systems, spreading codes are used for packet detection and synchronization purposes. Correlation techniques are used to identify and synchronize to its timing. In many instances, the spreading code sequence can be in the order of 1000 chips or more. Since the receiver must correlate through all possible delays, this process can result in unacceptable delays. [0003] To ameliorate this problem, a short spreading code with good aperiodic autocorrelation can be used for packet detection and synchronization purposes. One example is the IEEE 802.11 Wireless Local Area Network (WLAN) system, which uses a length 11 Barker code as a spreading sequence for the preamble and the header of a packet. The short length of the spreading sequence makes it easy for receivers to quickly detect the presence of a packet in the communication channel and to synchronize to its timing. [0004] In the case of a linear channel, for the purpose of receiver design, it is often desirable to estimate the impulse response of the communication channel. In the context of the WLAN, a multi-path linear channel is often utilized, and such communication channels require equalization for effective reception. Given an estimate of the impulse response of the communication channel, we can directly calculate equalizer coefficients through matrix computations, as opposed to the conventional adaptive algorithms. This is described in "Digital Communications," by John G. Proakis, 4th edition, August 15, 2000, which reference is hereby incorporated by reference herein. This allows equalizer coefficients to be computed in a digital signal processor (DSP) instead of in more expensive and less adaptable dedicated hardware implementing the adaptation algorithms. [0005] Unfortunately, because the spreading code used is short (e.g. on the order of 11 symbols) a straightforward correlation using the spreading code will produce a distorted estimate. What is needed is a simple, computationally efficient technique that can be used to compute substantially undistorted communication channel impulse response estimates, even when the received signal was chipped with a short spreading code. The present invention satisfies that need. SUMMARY OF THE INVENTION [0006] To address the requirements described above, the present invention discloses a method and apparatus for estimating a communication channel impulse response h(t). The method comprises the steps of generating com(t) = co(t + mNTc) for m = 0,l,A 9M by correlating a received signal r(t) with a spreading sequence St of length N9 wherein the received signal r(t) comprises a chip sequence c. applied to a communication channel characterizable by an impulse response h(t), and wherein the chip sequence cy is generated from a data sequence di spread by the spreading sequence 5,-; generating an estimated communication channel impulse response hM(t)%s a combination of com(i) and dm for ?n = 0,l,A ,M; and filtering the first estimated communication channel impulse response hM (t) to generate the estimated communication channel impulse response /z(f)with a filter / selected at least in part according to the spreading sequence Sg. The apparatus comprises a correlator for generating com(t) = co(t + mNTc) for w = 0,l,A ,M by correlating a received signal r(t) with a spreading sequence S§ of length N9 wherein the received signal r(f) comprises a chip sequence c. applied to a communication channel characterizable by an impulse response h(i), and wherein the chip sequence c. is generated from a data sequence di spread by the spreading sequence 5f; an estimator for generating an estimated communication channel impulse response hM (t) as a combination of com (t) and dm for m = 0,l,A 9M; and a filter / selected at least in part according to the spreading sequence Sg , the filter for filtering the first estimated communication channel impulse response hM(t) to generate the estimated communication channel impulse response h(t). [0007] The foregoing permits the impulse response h(t) of the communication channel to be accurately estimated, even with short chip codes. Non-intuitively, in the case of a time-limited channel impulse response, the present invention yields an estimate that can be made perfect in the limit of high signal-to-noise ratio (SNR). BRIEF DESCRIPTION OF THE DRAWINGS [0008] Referring now to the drawings in which like reference numbers represent corresponding parts throughout: [0009] FIG. 1 is a diagram of a transceiver system; [0010] FIG. 2 is a block diagram illustrating process steps that can be used to implement the present invention; [0011] FIG. 3 is a diagram of a transceiver system utilizing a filter/to improve the estimated communication channel impulse response; [0012] FIG. 4 is a diagram showing the response of the filter; [0013] FIG. 5 is a flowchart describing exemplary processing steps that can be used to improve the reconstruction of the value of the communication channel impulse response using super codes imposed on the portion of the data sequence; [0014] FIG. 6 is a diagram of a transceiver system utilizing super code to transmit sequences; [0015] FIG. 7 is a diagram presenting a correlator output using 11 symbol long Barker code; [0016] FIG. 8 is a diagram presenting a correlator output using Walsh codes as an input super code; [0017] FIG. 9 is a diagram presenting a correlator output after postprocessing with a filter/as described in FIG. 2 and FIG. 3; [0018] FIG. 10 is a diagram presenting a more detailed view of the main lobe peak, showing the estimate of the communication channel impulse response in the actual communications channel impulse response; and [0019] FIG. 11 is a diagram presenting one embodiment of a processor that can be used to practice the present invention. DETAILED DESCRIPTION OF PREFERRED EMBODIMENTS [0020] In the following description, reference is made to the accompanying drawings which form a part hereof, and which is shown, by way of illustration, several embodiments of the present invention. It is understood that other embodiments may be utilized and structural changes may be made without departing from the scope of the present invention. System Model [0021] FIG. 1 is a diagram of a transceiver system 100. Using signal spreader 103, a random data symbol sequence d. 102, comprising a series of data packets 128 (each of which may include a preamble 124 used by the receiver for identification purposes, as well as a data payload 126) is spread by a sequence 5-104 of length N:{Sn,0 Eq. (1) [0022] This spread chip sequence c; 106 is transmitted through a linear transmission channel 108 having a combined channel impulse response h(t). The transmitted signal is received by a receiver 112. The received waveform r(i) 114 is: Eq. (2) where ?i(t) 121 is an additive noise component. [0023] This formulation does not explicitly impose a causality requirement on h(t) 108. If explicit causality is desired, this can be accomplished by setting h{t) - 0, t Complex sequences could be easily accommodated if needed, but they are not common for synchronization purposes. [0024] The receiver 112 receives the transmitted signal, and correlates the received signal r(t) 114 with the known spreading sequence St 104 to identify the data as intended to be received by the receiver 112. Once the received signal r(t) 114 is received, the preamble can be examined to determine the address of the data and whether further processing is necessary. [0025] Such systems also use the received signal to estimate fee input response of the communication channel 108. This information is used to improve later detection and reception of signals from the transmitter 110. In circumstances where the spreading sequence St. 104 is relatively short, the data packet 128 must be detected quickly, and there is less data available to estimate the response of the communication channel 108. Conventional Detection and Synchronization [0026] For detection and synchronization purposes, the search for the spreading code is conventionally performed by correlating the received signal r(t) 114 with the spreading sequence. This is accomplished by the correlator 116. Although this correlation is typically done after sampling in the time domain, for notational simplicity, we do not perform the time domain discretization. The correlator 116 output co(t) 118 is given by: where !)(/) is the correlation between the chip sequence and the spreading sequence and we will refer to it as the chip correlation. lobes disappear. However, when the spreading sequence S. 104 is short, the sidelobes of the autocorrelation are not negligible and will cause significant distortion to the estimate of the communication channel impulse response h(t). Improved Channel Estimates for Short Spreading Sequences [0042] As is demonstrated below, the present invention improves the communication channel impulse response estimate by filtering the first estimated communication channel impulse response hM (t) to generate the estimated communication channel impulse response h(t)vnHi a filter/ selected at least in part according to the spreading sequence Ss. In particular, when the time span of the communication channel 108 is limited, a zero-forcing deconvolution can be used to improve the estimate. [0043] FIG. 2 is a block diagram illustrating process steps that can be used to implement the present invention. [0044] FIG. 3 is a diagram of a transceiver system 300 utilizing the filter / described above to filter the first estimated communication channel impulse response hM (t) to generate an improved estimate suitable for short spreading sequences Sf 104. [0045] Referring to FIG. 2 and FIG. 3, blocks 202 througji 208 recite steps that are used to generate com(t) IIS. A spread chip sequence c. 106 is generated from a data symbol sequence dt 102 and a spreading sequence 5,- 104 of length N, as shown in block 202. The chip sequence c. 106 is transmitted via a communication channel 108 as shown in block 204, and received as shown in block 206. The communication channel includes the transmitter 110 and the receiver 112. The received signal r(t) 114 is then correlated with the spreading sequence S. 104, by the correlator 116 to generate com (t) as shown in block 208. [0046] In block 210, an estimated communications channel impulse response hM (/) is generated by the estimator 120 as a combination of com(f)mA. dm for m = 0,l,K ,M This can be accomplished, for example using the relationship described in Eq. (24) above. [0052] Then, the filtered estimate ft,(or, in the earlier notation, hf(f)) is composed of an exact copy of h(h(t))9 plus some aliased versions of it in non-overlapping locations. So in this case h is resolvable from hf. [0053] Such a filter with length 21 + 1 can be designed with the simple zero- forcing criteria: Sf 104. L can be chosen such that the product LTC (the chip period Tc is known) is approximately equal to the time span (e.g. the approximate duration of the impulse response) of the channel 108. [0054] Note that the value A(n - i) is well defined ... it is a property of the spreading sequence S( 104, which is known apriori. [0055] As usual, the matrix structure of the linear equations is Toeplitz. By the design requirement of the spreading sequence Sg9 the matrix should be well conditioned. The filter coefficients can be computed offline given the spreading sequence and desired window width L. [0056] While the foregoing has been described in respect to non recursive filters, other filters, such as recursive filters may also be used. A recursive filter, for example, may provide perfect filtering of the sidelobes, but the result may not be the quell conditioned matrix, hence the solution may be more difficult to determine. In fact, any filter of length 2L +1 can be defined. Super Coded Transmit Sequences [0057] It has been shown that given h and with filtering, it is possible to recover the true channel impulse response for a time limited channel. However, in the foregoing [0062] If the condition that - /,N > (27V + L) I 12N>(2N + L) can be satisfied, h can be reconstructed free of aliasing interference, and by deconvolution (aforementioned filtering technique), h can be reconstructed as well. [0063] From the foregoing, it can be determined that a small supercode imposed on a portion of the data sequence can provide an alias free estimate of the communication channel impulse response when the channel response is time-limited. The only source of distortion from this estimate comes from the additive noise, which can be suppressed by the spreading gain times a factor of 2 (to account for the supercode). When the noise is low, such an approach is preferable over long integrations. [0064] For moderate values of Z, such code sequences can be easily embedded within a longer preamble to packet data, probably with multiple copies, without adversely affecting the spectrum properties of the transmission. In addition, when the signal to noise ratio (SNR) is low, traditional integration as outlined in the first half of this section can still be carried out on such a preamble to obtain a higher processing gain against the additive noise. [0065] FIG. 5 is a flowchart describing exemplary processing steps that can be used to improve the reconstruction of the value of the communication channel impulse response by using supercode imposed on a portion of the data sequence. [0066] FIG. 6 is a diagram of a transceiver system 600 utilizing super coded transmit sequences to generate an improved communication channel impulse response estimate suitable for short spreading sequences £,104. [0067] In block 502, a data sequence d{ 102 is generated. The data sequence^ 102 includes one or more data packets 128, each data packet having a preamble 124 including a constrained portion Cdi 602. The preamble 124, can be, for example, in the form of a pseudorandom code. [0068] The constrained portion Cdi 602 is associated with at least two codes, w0 and Wj. The codes w0 and w} are selected such that the correlation Acode{k) of the constrained portion Cd. 602 and at least one of the codes w0 and wx, is characterized by a maximum value at k = 0, and they value less than the maximum value at k * 0. [0069] Ideally, the correlation Acode(k) of the constrained portion Cdi 602 is an impulse, with Acode{k) equal to one at £ = 0, and equal at all other values for h However, because such correlation characteristics are typically not realizable, codes w0 and wx can be chosen to approximate this ideal. For example, codes w0 and wx can be chosen such that the correlation ^e(fc) of the constrained portion Cdi 602 and at least one of the codes wQ and wl9 is such that Acode(k) = l at & = 0and ^code(fc)«0for substantially all k * 0. Or, codes w0 and w^ can be chosen such that the correlation Acode(k) of the constrained portion Cdi 602 and at least one of the codes w0 and wl9is such that Acode(k) = 0 for 0 fc*0. [0070] In one embodiment, the constrained portion Cdi 602 comprises the pair of length two Walsh codes in the first sequence described above. Other embodiments are envisioned in which the codes are of another length (other than length two), or are codes other than a Walsh code. [0071] In block 504, a chip sequence Cj 106 is generated. The chip sequence c. 106 is generated by applying a spreading sequence Si 104 of length N and having a chip period Tc to the data sequence dx 102. [0072] This spread chip sequence c. 106 is transmitted through a linear transmission channel 108 having a combined channel impulse response h(t). The transmitted signal is received by a receiver 112. [0073] In block 506, the receiver 112 receives the transmitted signal, and correlates the received signal r(t) 114 with the known spreading sequence 5,104 to identify the data as intended to be received by the receiver 112. This is accomplished by generating com (t) = co(t + mNTc) for m = 0,1,A 9M, using techniques analogous to those which were described above. [0074] In block 508, an estimated communication channel impulse responsehM(t) is generated as a combination of the correlationcom(t) and the data sequence dm for m = 0,1,A 9M. adjustments were made for group delays introduced by correlation, filtering and windowing, therefore time coordinates should be treated in the relative sense. FlGs. 7-10 also do not include the effects of additive noise. [0092] FIG. 7 is a diagram presenting a correlator 116 output using a length 11 Barker code and conventional communication channel impulse response estimation techniques. The correlator 116 output shows a main lobe peak 702, and multiple spurious peaks 704. These spurious peaks 704 (which are 11 chips, or NTC seconds, apart due to the length 11 Barker code) are due to the repeated transmission of the short code S; 104, which are "aliased" back upon each other. If the length of the periodic spreading sequence 5-104 were longer, there would be fewer spurious peaks 704, and the peaks 704 would not overlap the main lobe peak 702 as much as is shown in FIG. 7. [0093] FIG. 8 is a diagram presenting a correlator 116 output using the Walsh codes in conjunction with the supercode technique described in FIG. 5. To generate this plot, the input data was constrained with two symbol-long Walsh codes wQ and w}9 and the output was processed by summing two successive outputs of the correlator 116 as shown in Eq. (36). For the 11 chips on either side of the main lobe peak 702, there is zero correlation, and many of the spurious correlator peaks 704 that were apparent in FIG. 7 are no longer evident. Note, however that since only six bits of the data sequence are constrained ...+,+,+,-,-,-..., some aliased versions of the main lobe peak 704 (labeled 802) are present (33 chips from the main lobe peak 702). However, since these aliased versions 802 are widely separated from the main lobe peak 702, an accurate estimate of the communications channel impulse response can be obtained. Note that a similar result can also be achieved without constraining the input sequence with the super code, but this would require integration over large number of symbols (e.g. M in Eq. (26) would be large). Note also that the main lobe peak 704 still includes minor peaks because the estimator 120 produces h, which is a smeared version of h. These undesirable components 804, caused by the autocorrelation of the spreading sequence 104, cannot be removed by constraining the data sequence. Instead, these undesirable components 804 can be removed by filtering as described with respect to FIG. 9 below. [0094] FIG. 9 is a diagram presenting a correlator 116 output shown in FIG. 8 after postprocessing with a filter/as described in FIGs. 2 and 3. Note that the sidelobes 802 shown in FIG. 8, have been pushed away from the main lobe peak 702, and some of the undesirable components 804 of the main lobe peak 702 have been filtered. Also note that the data indexing (the chips shown as the time axis) of FIG. 9 has changed relative to the data indexing of FIG. 8. As described above, this difference is an artifact of the software used to plot FIG. 7- FIG. 11 and is not associated with the applicant's invention. [0095] FIG. 10 is a diagram presenting a more detailed view of the main lobe peak 702, showing the estimate of the communication channel impulse response (indicated by the asterisks) and the actual communication channel impulse response. Note that the estimated communication channel impulse response follows that of the actual response very closely. Hardware Environment [0096] FIG. 11 and is a diagram illustrating an exemplary processor system 1102 that could be used in the implementation of selected elements of the present invention (including, for example, portions of the transmitter 110, the receiver 112, the correlator 116, the estimator 120, or the filter 302). [0097] The processor system 1102 comprises a processor 1104 and a memory 1106, such as random access memory (RAM). Generally, the processor system 1102 operates under control of an operating system 1108 stored in the memory 1106. Under control of the operating system 1108, the processor system 1102 accepts input data and commands and provides output data. Typically, the instructions for performing such operations are also embodied in an application program 1110, which is also stored in the memory 1106. The processor system 1102 may be embodied in a microprocessor, a desktop computer, or any similar processing device. [0098] Instructions implementing the operating system 1108, the application program 1110, and the compiler 1112 may be tangibly embodied in a computer-readable medium, e.g., data storage device 1124, which could include one or more fixed or removable data storage devices, such as a zip drive, floppy disc drive, hard drive, CD-ROM drive, tape drive, etc. Further, the operating system 1108 and the application program 1110 are comprised of instructions which, when read and executed by the computer 1102, causes the computer 1102 to perform the steps necessary to implement and/or use the present invention. Application program 1110 and/or operating instructions may also be tangibly embodied in memory 1106 and/or data communications devices 1130, thereby making an application program product or article of manufacture according to the invention. As such, the terms "article of manufacture," "program storage device" and "computer program product'" as used herein are intended to encompass a computer program accessible from any computer readable device or media. [0099] Those skilled in the art will recognize many modifications may be made to this configuration without departing from the scope of the present invention. For example, those skilled in the art will recognize that any combination of the above components, or any number of different components, peripherals, and other devices, may be used with the present invention. For example, an application-specific integrated circuit (ASIC) or a Field-Programmable Gate Array (FPGA) can be used to implement selected functions, including the correlator 116, and filtering functions can be performed by a general-purpose processor, as described above. Conclusion [0100] This concludes the description of the preferred embodiments of the present invention. The foregoing description of the preferred embodiment of the invention has been presented for the purposes of illustration and description. It is not intended to be exhaustive or to limit the invention to the precise form disclosed. Many modifications and variations are possible in light of the above teaching. It is intended that the scope of the invention be limited not by this detailed description, but rather by the claims appended hereto. The above specification, examples and data provide a complete description of the manufacture and use of the composition of the invention. Since many embodiments of the invention can be made without departing from the spirit and scope of the invention, the invention resides in the claims hereinafter appended. CLAIMS What is Claimed is: 1. A method of estimating a communication channel impulse response h(t), comprising the steps of: generating com(t) = co(t + mNTc) for m = 0,l,A,M by correlating a received signal r(i) with a spreading sequence Ss of length N, wherein the received signal r(0 comprises a chip sequence c. applied to a communication channel characterizable by an impulse response h(t), and wherein the chip sequence Cj is generated from a data sequence dt spread by the spreading sequence £,.and wherein Tc is the chip period of the chip sequence c.; generating an estimated communication channel impulse response hM(t)as a combination of com(t) and dm for m = O,1?A 9M; and filtering the first estimated communication channel impulse response hM (t) to generate the estimated communication channel impulse response h(t)vriih a filter/ selected at least in part according to the spreading sequence St. 2. The method of claim 1, wherein the filter/ is further selected at least in part according to an autocorrelation A(n) of the spreading sequence St. 3. The method of claim 2, wherein the filter/ is further selected at least in part according to the duration of the impulse response of the communication channel *(0- 4. The method of claim 2, wherein the filter/ is further selected at least in part according to a zero-forcing criteria wherein: /(/)is the impulse response of the filter/such that Af(n)is a convolution of A(n) and /(/); Sr 5. The method of claim 4, wherein: the parameter L is chosen such that a time duration of the impulse response of the communication channel h(t) is less than LTC. 6. The method of claim 4, wherein: the parameter L is chosen such that a time duration of the impulse response of the communication channel h{t)is approximately equal to LTC. 7. The method of claim 1, wherein N is less than 20. 8. The method of claim 1, wherein M= 0. 9. The method of claim 1, wherein the data sequence di includes a constrained portion Cds associated with at least two codes w09w}, wherein a correlation Acode(k) of the constrained portion Cdt with one of the codes M>0,WJ is characterized by a maximum value at k = 0 less than maximum values at k * 0. 10. The method of claim 9, wherein the step of generating an estimated communication channel impulse response hM (t) as a combination of com(t) and dm for m = 0,l?A yM comprises the step of computing 11. The method of claim 10, wherein M^2. 12. The method of claim 9, wherein the data sequence di includes a preamble having a pseudorandom code including the constrained portion of the data sequence dt. 13. The method of claim 9, wherein Acode(k) = 1 at k = 0 and AaHk(k) = 0 for substantially all k * 0. 14. The method of claim 9, wherein Acode(k) = 0 for 0 selected to minimize the correlation of the constrained portion Cd- with the one of the codes w0, M\ for substantially all k^O. 15. The method of claim 14? wherein 2/is a length of the constrained portion Cdt. 16. The method of claim 1, wherein Acode(k) = 1 at k = 0 and Acode(k) = Ofor substantially all k * 0. 17. The method of claim 1, wherein each of the two codes wo,w^ comprises two symbols. 18. The method of claim 1, wherein the each of the two codes w0, wx comprises no more than two symbols. 19. The method of claim 1, wherein the codes w09wx comprise Walsh codes. 20. An apparatus for estimating a communication channel impulse response h(f), comprising: means for generating com (t) = co(t + mNTc) for m - 0,1,A 9M by correlating a received signal r(t) with a spreading sequence Si of length AT, wherein the received signal r(0 comprises a chip sequence c. applied to a communication channel characterizable by an impulse response h(t), and wherein the chip sequence c. is generated from a data sequence d( spread by the spreading sequence Sf and wherein Tc is the chip period of the chip sequence c.; means for generating an estimated communication channel impulse response hM (r) as a combination of com (t) and dm for m = 0,1,A ,M; and a filter means /, selected at least in part according to the spreading sequence S{, the filter means for filtering the first estimated communication channel impulse response hM(t)io generate the estimated communication channel impulse response h(t) with 21. The apparatus of claim 20> wherein the filter means/ is further selected at least in part according to an autocorrelation A(n) of the spreading sequence 5-. 22. The apparatus of claim 21, wherein the filter means/ is further selected at least in part according to the duration of the impulse response of the communication channel h(i). 23. The apparatus of claim 21, wherein the filter means/ is further selected at least in part according to a zero-forcing criteria /(/)is the impulse response of the filter means / such that ^4/(n)is a convolution of A(n) and f(i); s,. 24. The apparatus of claim 23, wherein: the parameter L is chosen such that a time duration of the impulse response of the communication channel h(f) is less than LTC. 25. The apparatus of claim 23, wherein: the parameter L is chosen such that a time duration of the impulse response of the communication channel h(t) is approximately equal to LTC. 26. The apparatus of claim 20, wherein AT is less than 20. 27. The apparatus of claim 20, wherein M- 0. 28. The apparatus of claim 20, wherein the data sequence df includes a constrained portion Cd. associated with at least two codes w09wl9 wherein a correlation Acode(k) of the constrained portion Cdi with one of the codes wQ9wx is characterized by a maximum value at k = 0 less than maximum values at k * 0. 29. The apparatus of claim 28, wherein the means for generating an estimated communication channel impulse response hM(f)as a combination of com(t) and dm for m = 0,1,A ,Mcomprises means for computing hM{t)as 30. The apparatus of claim 29, wherein M=2. 31. The apparatus of claim 28, wherein the data sequence dt includes a preamble having a pseudorandom code including the constrained portion of the data sequence dr 32. The apparatus of claim 28, wherein Acode(k) = 1 at k = 0 and i4corfc (fc) = 0 for substantially all k * 0. 33. The apparatus of claim 28, wherein Acode(k) = 0 for 0 i/is selected to minimize the correlation of the constrained portion Q/,.with the one of the codes w09wx for substantially all k * 0. 34. The apparatus of claim 33, wherein 2J is a length of the constrained portion Cdi. 35. The apparatus of claim 20, wherein A^^k) = 1 at k = 0 and i4cwfc (&) = 0 for substantially all k * 0. 36. The apparatus of claim 20, wherein each of the two codes w0, M\ comprises two symbols. 37. The apparatus of claim 20, wherein the each of the two codes w0, Wj comprises no more than two symbols. 38. The apparatus of claim 20, wherein the codes W09M\ comprise Walsh codes. 39. An apparatus for estimating a communication channel impulse response h(t), comprising: a correlator generating com (t) = co(i -f mNTc) for m = 0,1,A ,M by correlating a received signal r{i) with a spreading sequence Si of length N9 wherein the received signal r(t) comprises a chip sequence Cj applied to a communication channel characterizable by an impulse response h(t), and wherein the chip sequence c. is generated from a data sequence di spread by the spreading sequence Sf and wherein Tc is the chip period of the chip sequence c}; an estimator for generating an estimated communication channel impulse response hM (t) as a combination of com if) and dm for m = 0,1,A ,M; and a filter/selected at least in part according to the spreading sequence S. , the filter for filtering the first estimated communication channel impulse response hM (t) to generate the estimated communication channel impulse response hit). 40. The apparatus of claim 39, wherein the filter/ is further selected at least in part according to an autocorrelation A(n) of the spreading sequence Si. 41. The apparatus of claim 40, wherein the filter/ is further selected at least in part according to the duration of the impulse response of the communication channel hit).

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