Title of Invention	APPARATUS AND METHOD FOR GENERATING A RECEPTION BEAM WEIGHT IN A MOBILE COMMUNICATION SYSTEM USING AN ADAPTIVE ANTENNA ARRAY SCHEME
Abstract	A method for generating a reception beam weight for generating a reception beam from a reception signal, the method comprising the steps of determining a first error value by using a Constant Modulus (CM) scheme at a timing point, and a second error value by using a Decision-Directed (DD) scheme different from the CM scheme at the timing point (419); determining a CM scheme application weight and a DD scheme application weight differently by comparing with the first error value and the second error value; generating a third error value using a scheme that combines the CM scheme to which the CM scheme application weight is applied and the DD scheme to which the DD scheme application weight is applied (421); determining a reception beam weight using the reception signal, the third error value, and an output signal generated by applying the reception beam to the reception signal, wherein the reception beam weight is used for generating the reception beam (423).

Title of Invention

APPARATUS AND METHOD FOR GENERATING A RECEPTION BEAM WEIGHT IN A MOBILE COMMUNICATION SYSTEM USING AN ADAPTIVE ANTENNA ARRAY SCHEME

Abstract

A method for generating a reception beam weight for generating a reception beam from a reception signal, the method comprising the steps of determining a first error value by using a Constant Modulus (CM) scheme at a timing point, and a second error value by using a Decision-Directed (DD) scheme different from the CM scheme at the timing point (419); determining a CM scheme application weight and a DD scheme application weight differently by comparing with the first error value and the second error value; generating a third error value using a scheme that combines the CM scheme to which the CM scheme application weight is applied and the DD scheme to which the DD scheme application weight is applied (421); determining a reception beam weight using the reception signal, the third error value, and an output signal generated by applying the reception beam to the reception signal, wherein the reception beam weight is used for generating the reception beam (423).

Full Text	BACKGROUND OF THE INVENTION 1. field of thc Invention The present invention relates generally to an apparatus and method for receiving data in a mobile communication system using an Adaptive Antenna Array (AAA) scheme, and in particular, to an apparatus and method for receiving data using an adaptive reception beam weight generation scheme. 2. Description of the Related Art A "next generation mobile communication system" has evolved into a packet service communication system that transmits burst packet data to a plurality of mobile stations (MSs). The packet service communication system is designed to transmit mass data. Such a packet service communication system has been developing for high-speed packet service. In this regard, the 3rd Generation Partnership Project (3GPP). a standardization organization for an asynchronous communication scheme, proposes a High Speed Downlink Packet Access (HSDPA) to provide the high-speed packet service, while the 3rd Generation Partnership Project 2 (3GPP2). a standardization organization for a synchronous communication scheme, proposes a 1x Evolution Data Only Voice (1x EV-DO V) to provide the high-speed packet service. Both the HSDPA and the lx EV-DO V intend to provide high-speed packet service for smooth transmission of Web Internet service, and in order to provide the high-speed packet service, a peak throughput and average throughput should be optimized for smooth transmission of the packet data as well as the circuit data. e.g.. voice service data. In order to support the high-speed transmission of packet data, a communication system employing the IISDPA (hereinafter referred to as an "IISDPA communication system") has newly introduced 3 kinds of data transmission schemes: an Adaptive Modulation and Coding (AMC) scheme: a Hybrid Automatic Retransmission Request (HARQ) scheme: and a Fast Cell Selection (FCS) scheme. The HSDPA communication system increases a data rate using the AMC. HARQ. and FCS sehemes. A communication system using the 1x EV-DO V (hereinafter referred to as a "1x EV-DO'V communication system") is another communication system for increasing a data rale. The lx EV-DO V communication system also increases a data rate to secure system performance. Aside from the new schemes such as AMC, HARQ and FCS. there is a Multiple Antenna scheme, which is another scheme for coping with the limitation in assigned bandwidth, i.e.. increasing a data rate. The Multiple Antenna scheme can overcome the limitation of bandwidth resource in a frequency domain because it utilizes a space domain. A communication system is constructed such that a plurality of MSs communicate with each other via one base station (BS). When the BS performs a high-speed data transmission to the MSs. a fading phenomenon occurs due to a characteristic of radio channels. In order to overcome the fading phenomenon, a Transmit Antenna Diversity scheme, which is a kind of the Multiple Antenna scheme, has been proposed. The Transmit Antenna Diversity scheme transmits signals using at least two transmission antennas to minimize a loss of transmission data due to the fading phenomenon, thereby increasing a data rale. Generally, in a wireless channel environment in a mobile communication system, unlike in a wired channel environment, a transmission signal is actually distorted due to several factors, such as multipath interference, shadowing, wave attenuation, lime-varying noise, interference, etc. Fading caused by the multipath interference is closely related to the mobility of a reflector or a user (or aMS). and actually, a mixture of a transmission signal and an interference signal is received. Therefore, the received signal suffers from severe distortion during its actual transmission, thereby reducing performance of the entire mobile communication system. The fading may result in the distortion in the amplitude and the phase of the received signal, preventing high-speed data communication in the wireless channel environment. Many studies are being conducted in order to resolve the fading. Accordingly, in order to transmit data at a high speed, the mobile communication system must minimize a loss caused by a characteristic of a mobile communication channel, such as fading, and interference of an individual user. A diversity scheme is used to prevent unstable communication due to the fading, and multiple antennas are used to implement a Space Diversity scheme. Transmit Antenna Diversity is popularly used as a scheme for efficiently resolving the fading phenomenon. The Transmit Antenna Diversity scheme receives a plurality of transmission signals that have experienced an independent fading phenomena in a wireless channel environment, thereby coping with distortion caused by the fading. The Transmit Antenna Diversity is classified into Time Diversity, frequency Diversity. Multipath Diversity, and Space Diversity. In other words, a mobile communication system must cope well with the fading phenomenon that severely affects communication performance, in order to perform the high-speed data communication. As indicated above, the fading phenomenon must be overcome because it reduces the amplitude of a received signal up to several dB to tens of dB. For example, a Code Division Multiple Access (CDMA) scheme utilizes a Rake receiver that can achieve diversity performance using a delay spread of the channel. The Rake receiver is a kind of a Receive Diversity scheme for receiving multipath signals. However, the Receive Diversity used in the Rake receiver is disadvantageous in that it cannot achieve a desired diversity gain when the delay spread of the channel is relatively small. The Time Diversity scheme efficiently copes with burst errors occurring in a wireless channel environment using interleaving and coding, and is generally used in a Doppler spread channel. Disadvantageously, however, the Time Diversity does not work well in a low-speed Doppler spread channel. The Space Diversity scheme is generally used in a channel with a low delay spread such as an indoor channel and a pedestrian channel, which is a low- speed Doppler spread channel. The Space Diversity scheme achieves a diversity gain using at least two antennas. In this scheme, when a signal transmitted via one antenna is attenuated due to fading, a signal transmitted via another antenna is received, thereby acquiring a diversity gain. The Space Diversity is classified into Receive Antenna Diversity using a plurality of reception antennas and Transmit Antenna Diversity using a plurality of transmission antennas. In the Rcceive-Adaptive Antenna Array (Rx-AAA) scheme, hy calculating a scalar product of an appropriate reception beam weight vector and a signal vector of a reception signal received via an antenna array comprised of a plurality of reception antennas, a signal received in a direction desired by a receiver is maximized and a signal received in a direction not desired by the receiver is minimized. Herein, the reception beam weight represents a weight for generating the reception beam generated by the receiver in applying the Rx-AAA scheme. As a result, the Rx-AAA scheme amplifies only a desired reception signal to a maximum level, thereby maintaining a high-quality call and increasing the entire system capacity and service coverage. Although the Rx-AAA scheme can be applied to both a frequency Division Multiple Access (FDMA) mobile communication system and a Time Division Multiple Access (TDMA) mobile communication system, it will be assumed herein that the Rx-AAA scheme is applied to a communication system using CDMA schemes (hereinafter referred to as a "CDMA communication system"). FIG. 1 is a block diagram illustrating a structure of a BS receiver in a conventional CDMA mobile communication system. Referring to FIG. 1. the US receiver is comprised of N reception antennas (Rx ANT) including a first reception antenna 111, a second reception antenna 121. and an Nth reception antenna 131. N radio frequency (RF) processors including a first RF processor 112. a second RF processor 122-...-and an Nth RF processor 132. being mapped to the corresponding reception antennas. N mullipath searchers including a first multipath searcher 113. a second multipath searcher 123- ...- and an Nth multipath searcher 133. being coupled to the corresponding RF processors. I. fingers including a first finger 140-1. a second linger 140-2- ...- and an Lth finger 140-1.. for processing L mullipath signals searched by the mullipath searchers, a mullipath combiner 150 for combining multipath signals output from the L fingers, a deinterleaver 160. and a decoder 170. Signals transmitted by transmitters in a plurality of MSs are received at the N reception antennas over a mullipath fading radio channel. The first reception antenna 111 outputs the received signal to the first RF processor 112. F.ach of the RF processors -includes an amplifier, a frequency converter, a filter, and an analog-to-digital (AD) converter, and processes an RF signal. The first RF processor 112 RF-processes a signal output from the first reception antenna 111 to convert the signal into a baseband digital signal, and outputs the baseband digital signal to the first multipath searcher 113. The first multipath searcher 113 separates I. multipath components from a signal output from the first RF processor 112. The separated L multipath components are output to the first finger 140-1 to the Lth finger 140-L. respectively. The first finger 140-1 to the Lth finger 140-L. being mapped to the L multiple paths on a one-to-one basis, process the I. multipath components. Because the L multiple paths arc considered for each of the signals received via the N reception antennas, signal processing must be performed on Nxl. signals, and among the NxL signals, signals on the same path are output to the same finger. Similarly, the second reception antenna 121 outputs the received signal to the second RF processor 122. The second RF processor 122 RF-processes a signal output from the second reception antenna 121 to convert the signal into a baseband digital signal, and outputs the baseband digital signal to the second multipath searcher 123. The second multipath searcher 123 separates 1.. multipath components from a signal output from the second RF processor 122. and the separated I, multipath components are output to the first finger 140-1 to the I. finger 140-L. respectively. In this same manner, the NTh reception antenna 131 outputs the received signal to the NTh RF processor 132. The NTh RF processor 132 RF-processes a signal output from the NTh reception antenna 131 to convert the signal into a baseband digital signal, and outputs the baseband digital signal to the NTh multipath searcher 133. The NTh multipath searcher 133 separates F multipath components from a signal output from the NTh RF processor 132. and the separated L multipath components are output to the first finger 140-1 to the LTh finger 140-L, respectively. Accordingly, among the L muliipalh signals for the signals received via the N reception antennas, the same multipath signals arc input to the same fingers, For example, first multipath signals from the first reception antenna 111 to the NTh reception antenna 131 are input to the first linger 140-1. In the same manner. LTh multipath signals from the first reception antenna III to the NTh reception antenna 131 are input to the LTh finger 140-L. The first finger 140-1 to the LTh finger 140-1. are different only in signals input thereto and output therefrom, and arc identical in structure and operation. Therefore, only the lirst finger 140-1 will be described for simplicity. The first finger 140-1 has N despreaders including a first despreader 141, a second despreader 142. . and an Nth despreader 143. being mapped to the N multipath searchers, a signal processor 144 for calculating a weight vector for generating a reception beam using signals received from the N despreaders. and a reception beam generator 145 for generating a reception beam using the weight vector calculated by the signal processor 144. A lirst multipath signal output from the lirst multipath searcher 113 is input to the lirst despreader 141. The lirst despreader 141 despreads the first multipath signal output from the lirst multipath searcher 113 with a predetermined despreading code, and outputs the despread multipath signal to the signal processor 144 and the reception beam generator 145. Here, the despreading process is called "temporal processing." Similarly, a first multipath signal output from the second multipath searcher 123 is input to the second despreader 142. The second despreader 142 despreads the lirst multipath signal output from the second multipath searcher 123 with a predetermined despreading code, and outputs the despread multipath signal to the signal processor 144 and the reception beam generator 145. Similarly, a first multipath signal output from the Nth multipath searcher 133 is input to the Nth despreader 143. The Nth despreader 143 despreads the first multipath signal output from the Nth multipath searcher 133 with a predetermined despreading code, and outputs the despread multipath signal to the signal processor 144 and the reception beam generator 145. The signal processor 144 receives the signals output from the first despreader 141 to the Nth despreader 143. and calculates a reception beam weight set W_k for generating a reception beam. A set of first multipath signals output from the first multipath searcher 113 to the Nth multipath searcher 133 will be defined as " X R." The first multipath signal set X R represents a set of first multipath signals received via the first reception antenna 111 to the Nth reception antenna 131 at a kth point, and the first multipath signals constituting the first multipath signal set X R are all vector signals. The reception beam weight set WR represents a set of reception beam weights to be applied to the first multipath signals received via the first reception antenna 111 to the Nth reception antenna 131 at the kth point, and the reception beam weights constituting the weight set WR are all vector signals. A set of signals determined by despreading all of the first multipath signals in the first multipath signal set X R will be defined as yR . The despread signal set YR. of the first multipath signals represents a set of signals determined by despreading the first multipath signals received via the first reception antenna 111 to the Nth reception antenna 131 at the kth point, and the despread signals constituting despread signal set yR of the first multipath signals are all vector signals. Herein, for the convenience of explanation, the term "set" will be omitted, and the underlined parameters represent sets of corresponding elements. Each of the first despreaders 141 to the Nth despreaders 143 despreads the first multipath signal XR with a predetermined despreading code, such that the reception power of a desired reception signal is greater than the reception power of an interference signal by a process gain. The despreading code is identical to the spreading code used in the transmitters of the MSs. As described above, the despread signal y of the first multipath signal XR is input to the signal processor 144. The signal processor 144 calculates a reception beam weight WR with the despread signal Y of the first multipath signal XR and outputs the reception beam weight IT. to the reception beam generator 145. As a result, the signal processor 144 calculates the reception beam weight WR including a total of N weight vectors applied to the first multipath signal XR. output from the first reception antenna 111 to the Nth reception antenna 131. with the despread signals yR of a total of N first multipath signals oulpul from the first reception antenna 111 to the Nth reception antenna 131. The reception beam generator 145 recehes the despread signals yR of a total of the N first multipath signals XR and a total of the N reception beam weight vectors WR. The reception beam generator 145 generates a reception beam with a total of the N reception beam weight vectors W. calculates a scalar product of the despread signal yR of the first multipath signal X and the reception beam weight WR corresponding to the reception beam, and outputs the result as an output zk of the first finger 140-1. The output zk of the first linger 140-1 can be expressed as shown in Equation (1). Zk - Wk H yk (1) In Equation (1). H denotes a Hermitian operator, i.e.. a conjugate- transpose. A set zk. of output signals zk from I. fingers in the BS receiver is finally input to the multipath combiner 150. Although only the first finger 140-1 has been described, the other fingers, the second linger 140-2 to the Lth finger 140-1., are to the same as the first finger 140-1 in operation. The multipath combiner 150 combines the signals oulpul from the first finger 140-1 to the Lth linger 140-1., and outputs the combined signal to the deinterleaver 160. The deinterleaver 160 deinterleaves the signal output from the multipath combiner 150 in a deinterleaving method corresponding to the interleaving method used in the transmitter, and outputs the deinterleaved signal to the decoder 170. The decoder 170 decodes the signal output from the deinterleaver 160 in a decoding method corresponding to the encoding method used in the transmitter, and outputs the decoded signal as final reception data. The signal processor 144 calculates a reception beam weight WR such that a Mean Square Error (MSH) of a signal received from a MS transmitter, desired to be received by a predetermined algorithm, becomes minimized. The reception beam generator 145 generates a reception beam using the reception beam weight WR, generated by the signal processor 144. The process of generating a reception beam such that the MSI:, is minimized is called "spatial processing." When the Rx-AAA scheme is used in a CDMA mobile communication system, temporal processing and spatial processing are simultaneously performed. The operation of simultaneously performing temporal processing and spatial processing is called "spatial-temporal processing.*' The signal processor 144 receives multipath signals despread for each linger in the above-described manner, and calculates a reception beam weight capable of maximizing a gain of the Rx-AAA scheme according to a predetermined algorithm. The signal processor 144 minimizes the MSE. Currently, a great deal of research is being conducted on a reception beam weight calculation algorithm for adaplively minimizing the MSL:1. However, the reception beam weight calculation algorithm for adaptively minimizing the MSE is an algorithm for reducing errors on the basis of a reference signal, and this algorithm supports a Constant Modulus (CM) scheme and a Decision-Directed (DD) scheme as a blind scheme, when there is no reference signal. further, the algorithm for reducing errors on the basis of a reference signal has trouble converging into a minimum MSI value desired by the system in an environment where a channel such as a fast fading channel suffers from a rapid change, or an environment where a high-order modulation scheme such as 16-ary quadrature amplitude modulation (16QAM) is used. liven though it converges into a particular MSE value, the minimum MSE value is set to a relatively large value. When the minimum MSE value is set to a relatively large value, a gain that occurs from using the Rx-AAA scheme is reduced. Therefore, this algorithm is not suitable for a high-speed data communication system. SUMMARY OF THE INVENTION It is, therefore, an object of the present invention to provide an apparatus and a method for receiving data using an Adaptive Antenna Array scheme in a mobile communication system. It is another object of the present invention to provide an apparatus and a method for receiving data using an adaptive reception beam weight generation scheme in a mobile communication system using an Adaptive Antenna Array scheme. It is further another object of the present invention to provide an apparatus and a method for generating a reception beam having a minimum error value in a mobile communication system using an Adaptive Antenna Array scheme. In accordance with a first aspect of the present invention, there is provided a method of generating a reception beam weight for generating a reception beam from a reception signal, the method comprisingdelermining a first error value by using a first scheme at a timing point, and a second error value by using a second scheme different from the first scheme at the liming point, determining a first scheme application weight according to a difference between the first error value and the second error value, and a second scheme application weight according to the difference between the first error value and the second error value, generating a third error value using a scheme that combines the first scheme to which the first scheme application weight is used and the second scheme to which the second scheme application weight is used, determining a reception beam weight using the reception signal, the third error value, and an output signal generated by using the reception beam to the reception signal, wherein the reception beam weight is used for generating the reception beam. In accordance with a second aspect of the present invention, there is provided an apparatus of generating a reception beam weight for generating a reception beam from a reception signal, the apparatus comprising an error value combiner of determining a first error value by using a first scheme at a liming point, and a second error value by using a second scheme different from the first scheme at the timing point, determining a first scheme application weight according to a difference between the first error value and the second error value, and a second scheme application weight according to the difference between the first error value and the second error value: generating a third error value using a scheme that combines the first scheme to which the first scheme application weight is used and the second scheme to which the second scheme application weight is used, a weight calculator of determining a reception beam weight using the reception signal, the third error value, and an output signal generated by using the reception beam to the reception signal, wherein the reception beam weight is used for generating the reception beam. BRIEF DESCRIPTION OF THE ACCOMPANYING DRAWINGS The above and other objects, features, and advantages of the present invention will become more apparent from the following detailed description when taken in conjunction with the accompanying drawings in which: FIG. 1 is a block diagram illustrating a base station receiver in a conventional CDMA mobile communication system: FIG. 2 is a graph illustrating a characteristic of sigmoid function used in an embodiment of the present invention: FIG. 3 is a block diagram illustrating a base station receiver according to an embodiment of the present invention: FIG. 4 is a flowchart illustrating a signal reception procedure by a base station receiver according to an embodiment of the present invention: FIG. 5 is a diagram illustrating Constant Modulus (CM) schemes in an OFDM mobile communication system: FIG. 6 is a diagram schematically illustrating Decision Directed (DD) schemes in an OFDM mobile communication system using Binary Phase Shift Keying (BPSK): FIG. 7 is a graph illustrating a characteristic curve for a general reception beam weight generation schemes and a reception beam weight generation scheme according to an embodiment of the present invention: FIG 8 is a graph illustrating a characteristic curve according to the number of reception antennas of a base station receiver for an reception beam weight generation scheme according to an embodiment of the present invention: and FIG. 9 is a block diagram illustrating a structure of an OFDM mobile communication system according to an embodiment of the present invention. DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT Several preferred embodiments of the present invention will now be described in detail herein below with re Terence to the annexed drawings. In the drawings, the same or similar elements are denoted by the same reference numerals even though they are depicted in different drawings. In the following description, a detailed description of known functions and configurations incorporated herein has been omitted for conciseness. Before a description of the present invention is given, a model of a reception signal received at a receiver of a base station (BS) will be considered. It is assumed that a receiver of the BS includes a receive-antenna array having a plurality of reception antennas (Rx ANTs), and the receive-antenna array is generally mounted only in the receiver of the BS considering its cost and size, and is not mounted in a receiver of a mobile station (MS). That is. it is assumed that the receiver of the MS includes only one reception antenna. Additionally, although the present invention is applicable to all of mobile communication systems using Frequency Division Multiple Access (FDMA). Time Division Multiple Access (TDMA). Code Division Multiple Access (CDMA), and Orthogonal Frequency Division Multiplexing (OFDM), the present invention will be described with reference to an OFDM mobile communication system. A signal transmitted from a transmitter of an mth MS existing in a cell serviced by the BS is expressed as shown in Equation (2). In Kquation (2), sm(t) denotes a transmission signal of an m11 MS. pm denotes transmission power of the mth MS. bm(t) denotes a user information bit sequence of the mth MS. and em(t) denotes a user spreading code sequence of the mth MS. having a chip period of Tc. The transmission signal transmitted from the MS transmitter is received at a receiver of the BS over a multipath vector channel. It is assumed that channel parameters of the multipath vector channel are relatively and continuously changed, compared with the bit period Tb. Therefore, it is assumed that the channel parameters of the multipath vector channel are constant for certain bit periods. A complex base band reception signal for a first multipath of an mth MS. received at a receiver of the BS. is expressed by liquation (3). It should be noted that the reception signal of liquation (3) represents a base band signal determined by down-converting a radio frequency (RF) signal received at the BS receiver. In Equation (3). denotes a set of complex base band reception signals received through a first multipath of the mth MS. αm1 denotes a fading attenuation applied to the first multipath of the mth MS. Φm1 denotes a phase transition applied to the first multipath of the mth MS. τm1 denotes a lime delay applied to the first multipath of the mth MS. and denotes a set of array responses (ARs) applied to the first multipath of the mth MS. Because the BS receiver includes a plurality of antennas, for example. N reception antennas, a signal transmitted by the mth MS is received at the BS receiver via the N reception antennas. Therefore, the number of signals received via the first multipath is N. and N complex base band reception signals received via the first multipath of the mth MS constitute a set of the reception signals. Herein, for the convenience of explanation, the term "set" will be omitted, and the underlined parameters represent sets of corresponding elements. When a current linear antenna array is used, the array response am1 is defined as shown in Equations (4). In Equation (4). 'd' denotes a distance between separated reception antennas. λ denotes a wavelength at a frequency band in use. N denotes the number of the reception antennas, and 0m1 denotes direction-of-arrival (DOA) applied to the first multipath of the mth MS. If it is assumed that the number of MSs existing in a cell serviced by the BS is M and there are L multiple paths for each of the M MSs. a reception signal received at the BS becomes the sum of transmission signals transmitted from the M MSs and additive white noise (AWN), as represented in liquation (5). In Equation (5). n(l) denotes the additive while noise added to the transmission signals transmitted from the M MSs. It is assumed that a signal the BS desires to receive in the reception signal of Hquation (5) is x11, x11 represents a signal a first MS has transmitted via a first multipath. Because it is assumed thai a signal the BS desires to receive is x11. all signals except the signal x11 are regarded as interference signals and noise. Therefore. liquation (5) can be rewritten as shown in liquation (6). In liquation (6), i(l) denotes an interference signal, which is defined in liquation (7). below. The first term of Equation (7) is a transmission signal of a MS that the BS desires to receive, but represents the inter-path interference (IPI) by other multiple paths lhat the BS does not desire to receive. The second term (7) represents the multiple access interference (MAI) by Further, the x(l) is desprcad with a desprcading code C1(1-τ11) previously set in a first finger (1-1) for a corresponding multipath in a corresponding channel card of the BS receiver, i.e.. a channel card (m-1) assigned to the first MS. and the desprcad signal y(l) is defined in Fquation (8) below. The despreading code C1(1-τ11) is identical to the despreading code c,(t-Tn) used in a BS transmitter during signal transmission. The BS includes a plurality of receivers as described in conjunction with FIG. 1. Hach of the receivers is called a "channel card." and one channel card is assigned to one MS. As described in connection with FIG. 1. the channel card includes as many lingers as the number of multiple paths, and the fingers are mapped to corresponding multipath signals on a one-to-one basis. When the signal y(l) is generated by despreading the pre-despread signal x(l) with the despreading code C1(1-τ11), the power of a signal component the BS receiver desires to receive from among the reception signals is amplified by a gain G according to a characteristic of a despreader. It is noted that although the power of a signal component the BS receiver desires to receive is amplified by a process gain G. the power of the signal components the BS receiver does not desire to receive is not changed at all. Therefore, a correlation matrix between a reception signal before despreading and a reception signal after despreading can be calculated. In order to calculate the correlation matrix between a reception signal before despreading and a reception signal after despreading. the reception signal x(l) before despreading is sampled at a kth point, which is equal to the sampling point of the reception signal y(l) after desprcading. The signal obtained by sampling the reception signal x(l) before despreading at the kth point is represented by Equation (9). In conclusion, in order to calculate a correlation matrix between a reception signal x(l) before despreading and a reception signal y(l) after despreading. it is assumed that the signal of liquation (9) is acquired by sampling the reception signal x(l) before despreading at the kth point, which is equal to the sampling point of the reception signal y(l) after despreading. and that the reception signal x(l) before despreading and the reception signal y(l) after despreading are stationary. A description will now be made of a Least Mean Square (LMS) scheme and a Minimum Mean Square Error (MMSE) scheme herein below. In the LMS scheme, a set of reception signals belbre despreading. including complex reception signals received via N reception antennas at a particular time. i.e.. complex reception signals x1 to xN received via a first reception antenna to an Nth reception antenna, will be defined as x - [X1,X2,...,XN]T. Here. "T" is an operator representing a transpose operation. In addition, a set of reception signals after despreading the complex reception signals X1, X2.... xN received via the N reception antennas will be defined as y=[y1,y2,... ,yN]T The reception signal y after despreading is determined by the sum of a signal component s the BS receiver desires to receive and a signal component u the BS receiver does not desire to receive, as represented by liquation (10). A set of complex reception beam weight values to be multiplied by the complex reception signals x1, x2,..., xN received via the N reception antennas, i.e.. complex reception beam weights w1 to wN to be multiplied by complex reception signals x1 to xN received via the first reception antenna to the Nth reception antenna, will be defined as An output signal z from lingers in a particular user card. i.e.. a channel card assigned to a particular MS. is determined by calculating a scalar product of the reception beam weight w and the reception signal y after despreading. as represented by Equation (11). In liquation (11). 'i' denotes the number of reception antennas. The output signal z can be classified into a signal component wII s the BS receiver desires to receive, and a signal component wII u the BS receiver does not desire to receive, using liquation (10) and liquation (11). The LMS scheme minimizes errors of a known reference signal and a reception signal, and particularly, minimizes a cost function J(w). as given below in equation (12). In liquation (12). 'J' denotes a cost function, and a reception beam weight value w for minimizing the cost function value J must be determined, further, in Hquation (12). ck denotes a difference, or an error, between a reception signal and a desired reception signal, and dk denotes the desired signal. In a reception beam algorithm using a non-blind scheme, a pilot signal is used as the desired signal dk by way of example. However, the present invention proposes a reception beam algorithm using a blind scheme, such that a detailed description of the reception beam algorithm using the non-blind scheme will be omitted. In liquation (12). the cost function J is a type of a second-order convex function. Therefore, in order to minimize the cost function J. the cost function J must be differentiated until its value becomes 0. A differentiated value of the cost function .) is shown below in Equation (13). However, it is difficult to acquire an optimal reception beam weight wII in an actual channel environment in a single process, and because the reception signal y after despreading is input at each point, a recursive formula of liquation (14) should be used in order to adaptive!}' or recursively acquire the optimal reception beam weight wII. In Hqualion (14), 'k' denotes a kth point. wR denotes a reception beam weight at the kth point, µ denotes a constant gain, and vR denotes a trace vector at the kth point. The trace vector v at the kth point represents a vector for converging a differentiated value of the cost function J to a minimum value, for example. 0. That is. liquation (14) shows a process of updating a value generated before or after a constant gain µ from a given reception beam weight wR to be used at a current point in a direction of the trace vector vR as a reception beam weight wR-1 to be used at the next point. The scheme for delecting a desired reception signal d(k). proposed in the present invention, is called a "blind scheme." Due to the use of the blind scheme, a received signal should be adaptively converged using a particular estimation value, and a below scheme is used for the adaptive convergence of the received signal. A combination mode blind scheme is used for detecting a desired received signal d(k). In this case, an error function is expressed as shown in liquation (15). In Equation (15). arc delected error values by applying Constant Modulus (CM) scheme and Decision Directed (DD) scheme to the received signal for an adaptive convergence of the received signal. The will be described herein below. In the present invention, as indicated above. are detected by applying combination of the CM scheme and the DD scheme to the received signal for an adaptive convergence of the received signal. That is. when the value is increased, a value is also increased. Consequently, an influence of the DD scheme is increased to a total error value. In Equation (15). g(x) is an s shape-function (sigmoid function). Accordingly, in a region with a large influence of the CM scheme, an influence of the DD scheme is decreased. In contrast, in a region with a large influence of the DD scheme, an influence of the CM scheme is decreased. In Hqualion (15). the error value ek is an error value by combining a value by applying a weight ak to the ekCM and a value by applying a weight P to the ekDD. Herein, the weight ak is a weight applied to the CM scheme as -CM scheme application weight, and the weight βk is a weight applied to the DD scheme as "DD scheme application weight'. Therefore, the error value ek is an error value detected by setting adaptively the weight ak and the weight P according to whether or not the error value of the received signal was converged. Additionally, the characteristic of sigmoid function g(x) will be described herein below. The FIG. 2 is a graph illustrating the characteristic of sigmoid function used in an embodiment of the present invention. Referring to FIG. 2. the characteristic of sigmoid function is changed according to value 'a'. When the value a increases, a shape of sigmoid function is closed to an 's' shape. When the value a is equal to 1 (a _ 1). the shape of sigmoid function is closed to a "straight line" shape. That is, when the CM scheme application weight ak is increased, the DD scheme application weight βk is decreased. However, when the CM scheme application weight αk is decreased, the DD scheme application weight βk is increased. If the error value ek calculated using the CM scheme is greater than the error value ek calculated using the DD scheme, the error value ek will be calculated by combining a weighted CM scheme and a more weighted DD scheme, compared with the weighted CM scheme. Herein, the weighted CM scheme is used to apply the CM scheme application weight αk to the CM scheme. The weighted DD scheme is used to apply the DD scheme application w eight βk to the DD scheme. The "more weighted DD scheme as compared with the weighted CM scheme" refers to the DD scheme application weight βk being greater than the CM scheme application weight αk. In the same manner, the "more weighted CM scheme as compared with the weighted DD scheme refers" to the CM scheme application weight a being greater than the DD scheme application weight βk. Additionally, if the error value ek calculated by using the CM scheme is equal or smaller than the error value ek calculated by using the DD scheme, the error value ek will be calculated by combining a weighted DD scheme and a more weighted CM scheme, compared with the weighted DI) scheme. A constant modulus (CM) scheme, which is used for adaptive convergence of the received signal, is generally used in a blind equalizer and also used for a generation algorithm. When the CM scheme proposed by Godard is used, the cost function J is expressed as shown in Equation (16) below. In liquation (16). 'p' denotes a particular positive integer, and Rp denotes a Godard modulus. The Godard modulus Rp is defined as shown in liquation (17). Because the current OFDM mobile communication system generally uses a high-order modulation scheme, which is higher in order than quadrature phase shift keying (QPSK) modulation, the cost function .1 is separated into a real part and an imaginary part as shown in liquation (18). The cost function .1 is separated into a real part and an imaginary part because transmission reception signals in the high-order modulation scheme have a real part and an imaginary part. It is assumed herein that the present invention uses the FMS scheme and the MMSF. scheme, and p-2. Therefore. d(k)=R2.R+jR2.1. In addition, it is assumed that a cost function value J at an initial point, i.e.. a k=0 point, is 0 (J=0). FIG. 5 is a diagram illustrating a CM scheme in an OFDM mobile communication system. Referring to FIG. 5. a CM scheme for p=2. d(k)- R2.R-jR2.1, and J-0 at a point with k-0. That is, if a value R2 is determined by liquation (18). a circle is generated on a coordinate surface. Then, a received signal is determined as a point where an extension line drawn from the origin meets the circle. In FIG. 5. received zk is projected as a circle. Above, the convergence step has been described. Herein below, a stabilizing step for acquiring the d(k) will be described. If MSE is converged into a predetermined value through the convergence step, a change occurs from the convergence step to the stabilization step where calculation of Equation 19 is performed. A process where a change occurs from the convergence step to the stabilization step as the MSF. is converged into a predetermined value will be described later on. In the stabilization step, like in the convergence step, a real part and an imaginary part are separately calculated. In liquation (19). Pr denotes a received signal is projected as a signal most approximating the desired reception signal d(k) by a DD scheme. The DD scheme is a scheme for reflecting the d(k) as a decision value most approximating the received signal. FIG. 6 is a diagram illustrating a DD scheme in an OFDM mobile communication system using Binary Phase Shift Keying (BPSK). Referring to FIG. 6, because it is assumed that the 01 DM mobile communication system uses BPSK. if a reception signal is (1.2. -0.2) in an I-Q domain, the desired reception signal d(k) is projected as the largest approximate value of 1 after calculating a distance from - 1 and -1. FIG. 3 is a block diagram illustrating a BS receiver according to a first embodiment of the present invention. While describing FIG. 3. it should be noted that a BS receiver according to the First embodiment of the present invention is similar in structure to the BS receiver described in connection with FIG. 1. but different in a method for determining a reception beam weight by a signal processor. For simplicity, only the elements directly related to the present invention in the BS receiver will be described with reference to FIG. 3. Further, the first embodiment of the present invention corresponds to an embodiment where the LMS scheme is used. Referring to FIG. 3. when a reception signal x, at a timing point k is received, a despreader 310 despreads the reception signal xk, using a predetermined despreading code, and outputs the despread reception signal y to a signal processor 330 and a reception beam generator 320. The signal processor 330 includes a weight calculator 331. a memory 333. and an error value combiner 335. For simplicity. FIG. 3 will be described with reference to only the first finger 140-1 in the BS receiver of FIG. 1. Therefore, the desprcader 310 of FIG. 3 is substantially identical in operation to the N despreaders of the first despreader 141 to the Nth desprcader 143 in the first finger 140-1. The error value combiner 335 inputs the despread reception signal y . and combines an error value ek by using the CM scheme and the DD scheme. The weight calculator 331 in the signal processor 330 calculates a reception beam weight wR by receiving the combined error value ek . the despread reception signal yk a predetermined constant gain µ. and an initial reception beam weight w0. and outputs the calculated reception beam weight to the memory 333. The memory 333 buffers the reception beam weight wR calculated by the weight calculator 331, and the weight calculator 331 uses the reception beam weight wR stored in the memory 333 when updating the reception beam weight wR. That is. the weight calculator 331 updates a reception beam weight wR. at the next timing point k+1 using the reception beam weight wR calculated at the timing point k. FIG. 4 is a flowchart illustrating a signal reception procedure by a BS receiver according to an embodiment of the present invention. Referring to FIG 4. in step 411. a BS receiver establishes an initial reception beam weight it, . and a constant gain µ. In step 413. the BS receiver determines if the communication has ended. If it is determined that the communication has ended, the BS receiver ends the ongoing procedure. If it is determined in step 413 that the eommunieation has not ended, the BS receiver proceeds to step 415. In step 415, the BS receiver receives a despread signal yk for the reception signal x. In step 417. the BS receiver calculates a set zk of signals zk output from respective fingers of the BS receiver using the despread signal yk and a reception beam weight The zR represents a set of finger output signals generated using a reception beam generated using the reception beam weight wR. In step 419. the BS receiver calculates an error value ek to decrease an error between the reception signal xk and a desired reception signal . In step 421. the BS receiver calculates a differentiated value of a cost function using the despread signal y. and the error function ek (VJ(wR) = -2ekyk). In step 423. the BS receiver calculates a reception beam coefficient, or a reception beam weight In step 425. the BS receiver maintains the calculated reception beam weight wR. In step 427. the BS receiver delays by a predetermined unit lime. The predetermined unit time is delayed in order to use a value determined at a kth snap for a (k-1)th snap. i.e.. to take a slate transition delay into consideration. In step 429. the BS receiver increases the k by 1. i.e.. transitions from the current timing point k to the next timing point k+1. and then returns to step 413. FIG. 7 is a graph illustrating a characteristic curve for a general reception beam weight generation schemes and an reception beam weight generation scheme according to an embodiment of the present invention. Referring to FIG. 7. it is noted that an MSF value (y-axis) compared to a number of iterations (x-axis) for the reception beam weight generation scheme according to the present invention 703 is converged into a lower value, compared with an MSF value against the number of iterations for the conventional reception beam weight generation scheme 701, e.g.. a CM scheme. That the MSF value is converged into a less value means that a reception beam can be correctly generated, making it possible to correctly receive only a desired reception signal. FIG. 8 is a graph illustrating a characteristic curve according to the number of reception antennas of a BS receiver tor an adaptive reception beam weight generation scheme according to the embodiment of the present invention. Referring to FIG. 8. there is illustrated a radiation pattern for a BS receiver having 6 reception antennas and a BS receiver having 10 reception antennas. For example, if it is assumed that a particular BS is located at 57°. it is noted that compared with the BS receiver having 6 reception antennas, the BS receiver having 10 reception antennas has a normalized antenna gain of about 0.2. and can more correctly generate a reception beam. As a result, in terms of capacity of an OFDM mobile communication system, an increase in the number of the reception antennas causes an increase in the amplitude of the reception signals enabling a correct communication, thereby contributing to an increase in system capacity. FIG. 9 is a block diagram illustrating a structure of an OFDM mobile communication system according to an embodiment of the present invention. Referring to FIG. 9. the OFDM communication system includes a transmitter, i.e.. an MS transmitter 900. and a receiver, i.e.. a BS receiver 950. The MS transmitter 900 includes a symbol mapper 911, a serial-to-parallel (or S P) converter 913. a pilot pattern inserter 915. an inverse fast Fourier transform (IFFT) unit 917. a parallel-to-serial (or P/S) converter 919. a guard interval inserter 921. a digital-to- analog (D/A) converter 923. and a radio frequency (RF) processor 925. When there are information data bits to be transmitted, the information data bits are input to the symbol mapper 911. The symbol mapper 911 modulates the input information data bits in a predetermined modulation scheme for symbol mapping, and outputs the symbol-mapped data bits to the serial-to-parallel converter 913. Here, quadrature phase shift keying (QPSK) or 16-ary quadrature amplitude modulation (I6QAM) can be used as the modulation scheme. The serial-to-parallel converter 913 parallel-converts serial modulation symbols output from the symbol mapper 911. and outputs the parallel-converted modulation symbols to the pilot pattern inserter 915. The pilot pattern inserter 915 inserts pilot patterns in the parallel-converted modulation symbols output from the serial-to-parallel converter 913. and then outputs the pilot pattern-inserted modulation symbols to the IFFT unit 917. The IFFT unit 917 performs N-point IFFT on the signals output from the pilot pattern inserter 915. and outputs the resultant signals to the parallel-to-serial converter 919. The parallel-to-serial converter 919 serial-converts the signals output form the IFFT unit 917. and outputs the serial-convened signals to the guard interval inserter 921. The guard interval inserter 921 receives the signal output from the parallel-to-serial converter 919, inserts a guard interval therein, and outputs the guard interval-inserted signal to the digilal-to-analog converter 923. The guard interval is inserted to remove interference between a previous OFDM symbol transmitted at a previous OFDM symbol time and a current OFDM symbol to be transmitted at a current OFDM symbol time in an 01 DM communication system. For the guard interval, a cyclic prefix method or a cyclic postfix method is used. In the cyclic prefix method, a predetermined number of last samples of an OFDM symbol in a lime domain are copied and inserted into a valid OFDM symbol. In the cyclic postfix method, a predetermined number of first samples of an OFDM symbol in a lime domain arc copied and inserted into a valid OFDM symbol. The digilal-to-analog converter 923 analog-converts the signal output from the guard interval inserter 921. and outputs the analog-converted signal to the RF processor 925. The RF processor 925. including a tiller and a front-end unit (not shown). RF-proccsses the signal output from the digital-to-analog converter 923 such that the signal can be transmitted via an antenna. The BS receiver 950 includes an RF processor 951. an analog-to-digital (or A/D) converter 953. a reception beam generator 955. a signal processor 957. a guard interval remover 959. a serial-to-parallel (or SP) converter 961. a fast Fourier transform (FFT) unit 963. an equalizer 965. a pilot symbol extractor 967. a synchronization & channel estimation unit 969. a parallel-to-serial (or P S) converter 971. and a symbol demapper 973. The signals transmitted by the MS transmitter 900 are received via reception antennas of the BS receiver 950. The received signals experience a multipath channel and have a noise component. The signals received via the reception antennas are input to the RF processor 951. which down-converts the signals received via the reception antennas into an intermediate frequency (IF) signal, and outputs the IF signal to the analog-to-digital converter 953. The analog-to-digital converter 953 digital-converts an analog signal output from the RF processor 951. and outputs the digital-converted signal to the reception beam generator 955 and the signal processor 957. Operations of the reception beam generator 955 and the signal processor 957 have been described above with reference to the first and second embodiments of the present invention. Therefore, a detailed description thereof will not be given again. The signal output from the reception beam generator 955 is input to the guard interval remover 959. The guard interval remover 959 removes a guard interval from the signal output from the reception beam generator 955. and outputs the resultant signal to the serial-to-parallel converter 961. The serial-to- parallel converter 961 parallel-converts the serial signal output from the guard interval remover 959. and outputs the resultant signal to the IF 1 unit 963. The FFT unit 963 performs N-point FFT on the signal output from the serial-to- parallel converter 961. and outputs the resultant signal to the equalizer 965 and the pilot symbol extractor 967. The equalizer 965 performs channel equalization on the signal output from the FFT unit 963. and outputs a resultant signal to the parallel-to-serial converter 971. The parallel-to-serial converter 971 serial-converts the parallel signal output from the equalizer 965. and outputs a resultant signal to the symbol demapper 973. The symbol demapper 973 demodulates the signal output from the parallel-to-serial converter 971 using a demodulation scheme corresponding to the modulation scheme used in the MS transmitter 900. and outputs a resultant signal as received information data bits. The signal output from the FFT unit 963 is input to the pilot symbol extractor 967. and the pilot symbol extractor 967 extracts pilot symbols from the signal output from the FFT unit 963. and outputs the extracted pilot symbols to the synchronization & channel estimation unit 969. The synchronization & channel estimation unit 969 synchronizes and channel estimates the pilot symbols output from the pilot symbol extractor 967, and outputs the result to the equalizer 965. As is understood from the foregoing description, the mobile communication system according to the present invention generates a weight using an adaptive reception beam weight generation scheme combined a CM scheme and a DD scheme, thereby making it possible to rapidly generate a reception beam weight with a minimum error value. Therefore, it is possible to generate a correct reception beam, and the correct reception of a reception beam enables a receiver to correctly receive only a desired signal, thereby improving system performance. While the present invention has been shown and described with reference to certain preferred embodiments thereof, it will be understood by those skilled in the art that various changes in form and details may be made therein without departing from the spirit and scope of the invention as defined by the appended claims. WE CLAIM: 1. A method for generating a reception beam weight for generating a reception beam from a reception signal, the method comprising the steps of: determining a first error value by using a Constant Modulus (CM) scheme at a timing point, and a second error value by using a Decision- Directed (DD) scheme different from the CM scheme at the timing point (419); determining a CM scheme application weight and a DD scheme application weight differently by comparing with the first error value and the second error value; generating a third error value using a scheme that combines the CM scheme to which the CM scheme application weight is applied and the DD scheme to which the DD scheme application weight is applied (421); determining a reception beam weight using the reception signal, the third error value, and an output signal generated by applying the reception beam to the reception signal, wherein the reception beam weight is used for generating the reception beam (423). 2. The method as claimed in claim 1, wherein the CM scheme application weight and the DD application scheme application weight are determined using sigmoid function. 3. The method as claimed in claim 1, wherein the DD scheme application weight is determined as a value greater than the CM scheme application weight, if the first error value is greater than the second error value. 4. The method as claimed in claim 1, wherein the CM scheme application weight is determined as a value greater than the DD scheme application weight, if the first error value is not greater than the second error value. 5. The method as claimed in claim 1, wherein each of the first and the second the error values is a value representative of a difference between a desired reception signal and the output signal. 6. The method as claimed in claim 5, wherein each of the first and the second the error values is a mean square error (MSE) value. 7. An apparatus for generating a reception beam weight for generating a reception beam from a reception signal, the apparatus comprising : an error value combiner (335) of determining a first error value by using a Constant Modulus (CM) scheme at a timing point, and a second error value by using a Decision Directed (DD) scheme different from the CM scheme at the timing point, determining a CM scheme application weight and a DD scheme application weight differently by comparing with the first error value and the second error value; generating a third error value using a scheme that combines the CM scheme application weight is applied and the DD scheme to which the DD scheme application weight is applied ; a weight calculator (331) of determining a reception beam weight using the reception signal, the third error value, and an output signal generated by applying the reception beam to the reception signal, wherein the reception beam weight is used for generating the reception beam. 8. The apparatus as claimed in claim 7, wherein the error value combiner comprises means for determining the CM scheme application weight and the DD application scheme application weight using sigmoid function. 9. The apparatus as claimed in claim 7, wherein the error value combiner comprises means for determining the DD scheme application weight greater than the CM scheme application weight, if the first error value is greater than the second error value. 10. The apparatus as claimed in claim 7, wherein the error value combiner comprises means for determining the CM scheme application weight greater than the DD scheme application weight, if the first error value is not greater than the second error value. 11. The apparatus as claimed in claim 7, wherein each of the first and the second the error values is a value representative of a difference between a desired reception signal and the output signal. 12. The apparatus as claimed in claim 1 1, wherein each of the first and the second the error values is a mean square error (MSE) value. A method for generating a reception beam weight for generating a reception beam from a reception signal, the method comprising the steps of determining a first error value by using a Constant Modulus (CM) scheme at a timing point, and a second error value by using a Decision-Directed (DD) scheme different from the CM scheme at the timing point (419); determining a CM scheme application weight and a DD scheme application weight differently by comparing with the first error value and the second error value; generating a third error value using a scheme that combines the CM scheme to which the CM scheme application weight is applied and the DD scheme to which the DD scheme application weight is applied (421); determining a reception beam weight using the reception signal, the third error value, and an output signal generated by applying the reception beam to the reception signal, wherein the reception beam weight is used for generating the reception beam (423).

Full Text

BACKGROUND OF THE INVENTION
1. field of thc Invention
The present invention relates generally to an apparatus and method for
receiving data in a mobile communication system using an Adaptive Antenna
Array (AAA) scheme, and in particular, to an apparatus and method for receiving
data using an adaptive reception beam weight generation scheme.
2. Description of the Related Art
A "next generation mobile communication system" has evolved into a
packet service communication system that transmits burst packet data to a
plurality of mobile stations (MSs). The packet service communication system is
designed to transmit mass data. Such a packet service communication system has
been developing for high-speed packet service. In this regard, the 3rd Generation
Partnership Project (3GPP). a standardization organization for an asynchronous
communication scheme, proposes a High Speed Downlink Packet Access
(HSDPA) to provide the high-speed packet service, while the 3rd Generation
Partnership Project 2 (3GPP2). a standardization organization for a synchronous
communication scheme, proposes a 1x Evolution Data Only Voice (1x EV-DO V)
to provide the high-speed packet service. Both the HSDPA and the lx EV-DO V
intend to provide high-speed packet service for smooth transmission of
Web Internet service, and in order to provide the high-speed packet service, a
peak throughput and average throughput should be optimized for smooth
transmission of the packet data as well as the circuit data. e.g.. voice service data.
In order to support the high-speed transmission of packet data, a
communication system employing the IISDPA (hereinafter referred to as an
"IISDPA communication system") has newly introduced 3 kinds of data
transmission schemes: an Adaptive Modulation and Coding (AMC) scheme: a
Hybrid Automatic Retransmission Request (HARQ) scheme: and a Fast Cell
Selection (FCS) scheme. The HSDPA communication system increases a data rate

using the AMC. HARQ. and FCS sehemes.
A communication system using the 1x EV-DO V (hereinafter referred to
as a "1x EV-DO'V communication system") is another communication system for
increasing a data rale. The lx EV-DO V communication system also increases a
data rate to secure system performance. Aside from the new schemes such as
AMC, HARQ and FCS. there is a Multiple Antenna scheme, which is another
scheme for coping with the limitation in assigned bandwidth, i.e.. increasing a
data rate. The Multiple Antenna scheme can overcome the limitation of
bandwidth resource in a frequency domain because it utilizes a space domain.
A communication system is constructed such that a plurality of MSs
communicate with each other via one base station (BS). When the BS performs a
high-speed data transmission to the MSs. a fading phenomenon occurs due to a
characteristic of radio channels. In order to overcome the fading phenomenon, a
Transmit Antenna Diversity scheme, which is a kind of the Multiple Antenna
scheme, has been proposed. The Transmit Antenna Diversity scheme transmits
signals using at least two transmission antennas to minimize a loss of
transmission data due to the fading phenomenon, thereby increasing a data rale.
Generally, in a wireless channel environment in a mobile communication
system, unlike in a wired channel environment, a transmission signal is actually
distorted due to several factors, such as multipath interference, shadowing, wave
attenuation, lime-varying noise, interference, etc. Fading caused by the multipath
interference is closely related to the mobility of a reflector or a user (or aMS). and
actually, a mixture of a transmission signal and an interference signal is received.
Therefore, the received signal suffers from severe distortion during its actual
transmission, thereby reducing performance of the entire mobile communication
system. The fading may result in the distortion in the amplitude and the phase of
the received signal, preventing high-speed data communication in the wireless
channel environment. Many studies are being conducted in order to resolve the
fading. Accordingly, in order to transmit data at a high speed, the mobile
communication system must minimize a loss caused by a characteristic of a
mobile communication channel, such as fading, and interference of an individual
user. A diversity scheme is used to prevent unstable communication due to the

fading, and multiple antennas are used to implement a Space Diversity scheme.
Transmit Antenna Diversity is popularly used as a scheme for efficiently
resolving the fading phenomenon. The Transmit Antenna Diversity scheme
receives a plurality of transmission signals that have experienced an independent
fading phenomena in a wireless channel environment, thereby coping with
distortion caused by the fading. The Transmit Antenna Diversity is classified into
Time Diversity, frequency Diversity. Multipath Diversity, and Space Diversity. In
other words, a mobile communication system must cope well with the fading
phenomenon that severely affects communication performance, in order to
perform the high-speed data communication.
As indicated above, the fading phenomenon must be overcome because it
reduces the amplitude of a received signal up to several dB to tens of dB. For
example, a Code Division Multiple Access (CDMA) scheme utilizes a Rake
receiver that can achieve diversity performance using a delay spread of the
channel. The Rake receiver is a kind of a Receive Diversity scheme for receiving
multipath signals. However, the Receive Diversity used in the Rake receiver is
disadvantageous in that it cannot achieve a desired diversity gain when the delay
spread of the channel is relatively small.
The Time Diversity scheme efficiently copes with burst errors occurring
in a wireless channel environment using interleaving and coding, and is generally
used in a Doppler spread channel. Disadvantageously, however, the Time
Diversity does not work well in a low-speed Doppler spread channel.
The Space Diversity scheme is generally used in a channel with a low
delay spread such as an indoor channel and a pedestrian channel, which is a low-
speed Doppler spread channel. The Space Diversity scheme achieves a diversity
gain using at least two antennas. In this scheme, when a signal transmitted via one
antenna is attenuated due to fading, a signal transmitted via another antenna is
received, thereby acquiring a diversity gain. The Space Diversity is classified into
Receive Antenna Diversity using a plurality of reception antennas and Transmit
Antenna Diversity using a plurality of transmission antennas.

In the Rcceive-Adaptive Antenna Array (Rx-AAA) scheme, hy
calculating a scalar product of an appropriate reception beam weight vector and a
signal vector of a reception signal received via an antenna array comprised of a
plurality of reception antennas, a signal received in a direction desired by a
receiver is maximized and a signal received in a direction not desired by the
receiver is minimized. Herein, the reception beam weight represents a weight for
generating the reception beam generated by the receiver in applying the Rx-AAA
scheme. As a result, the Rx-AAA scheme amplifies only a desired reception
signal to a maximum level, thereby maintaining a high-quality call and increasing
the entire system capacity and service coverage.
Although the Rx-AAA scheme can be applied to both a frequency
Division Multiple Access (FDMA) mobile communication system and a Time
Division Multiple Access (TDMA) mobile communication system, it will be
assumed herein that the Rx-AAA scheme is applied to a communication system
using CDMA schemes (hereinafter referred to as a "CDMA communication
system").
FIG. 1 is a block diagram illustrating a structure of a BS receiver in a
conventional CDMA mobile communication system. Referring to FIG. 1. the US
receiver is comprised of N reception antennas (Rx ANT) including a first
reception antenna 111, a second reception antenna 121. and an Nth reception
antenna 131. N radio frequency (RF) processors including a first RF processor
112. a second RF processor 122-...-and an Nth RF processor 132. being mapped
to the corresponding reception antennas. N mullipath searchers including a first
multipath searcher 113. a second multipath searcher 123- ...- and an Nth multipath
searcher 133. being coupled to the corresponding RF processors. I. fingers
including a first finger 140-1. a second linger 140-2- ...- and an Lth finger 140-1..
for processing L mullipath signals searched by the mullipath searchers, a
mullipath combiner 150 for combining multipath signals output from the L
fingers, a deinterleaver 160. and a decoder 170.
Signals transmitted by transmitters in a plurality of MSs are received at
the N reception antennas over a mullipath fading radio channel. The first
reception antenna 111 outputs the received signal to the first RF processor 112.

F.ach of the RF processors -includes an amplifier, a frequency converter, a filter,
and an analog-to-digital (AD) converter, and processes an RF signal. The first RF
processor 112 RF-processes a signal output from the first reception antenna 111 to
convert the signal into a baseband digital signal, and outputs the baseband digital
signal to the first multipath searcher 113. The first multipath searcher 113
separates I. multipath components from a signal output from the first RF
processor 112. The separated L multipath components are output to the first finger
140-1 to the Lth finger 140-L. respectively.
The first finger 140-1 to the Lth finger 140-L. being mapped to the L
multiple paths on a one-to-one basis, process the I. multipath components.
Because the L multiple paths arc considered for each of the signals received via
the N reception antennas, signal processing must be performed on Nxl. signals,
and among the NxL signals, signals on the same path are output to the same
finger.
Similarly, the second reception antenna 121 outputs the received signal to
the second RF processor 122. The second RF processor 122 RF-processes a signal
output from the second reception antenna 121 to convert the signal into a
baseband digital signal, and outputs the baseband digital signal to the second
multipath searcher 123. The second multipath searcher 123 separates 1.. multipath
components from a signal output from the second RF processor 122. and the
separated I, multipath components are output to the first finger 140-1 to the I.
finger 140-L. respectively.
In this same manner, the NTh reception antenna 131 outputs the received
signal to the NTh RF processor 132. The NTh RF processor 132 RF-processes a
signal output from the NTh reception antenna 131 to convert the signal into a
baseband digital signal, and outputs the baseband digital signal to the NTh
multipath searcher 133. The NTh multipath searcher 133 separates F multipath
components from a signal output from the NTh RF processor 132. and the
separated L multipath components are output to the first finger 140-1 to the LTh
finger 140-L, respectively.
Accordingly, among the L muliipalh signals for the signals received via

the N reception antennas, the same multipath signals arc input to the same fingers,
For example, first multipath signals from the first reception antenna 111 to the NTh
reception antenna 131 are input to the first linger 140-1. In the same manner. LTh
multipath signals from the first reception antenna III to the NTh reception antenna
131 are input to the LTh finger 140-L. The first finger 140-1 to the LTh finger 140-1.
are different only in signals input thereto and output therefrom, and arc identical
in structure and operation. Therefore, only the lirst finger 140-1 will be described
for simplicity.
The first finger 140-1 has N despreaders including a first despreader 141,
a second despreader 142. . and an Nth despreader 143. being mapped to the N
multipath searchers, a signal processor 144 for calculating a weight vector for
generating a reception beam using signals received from the N despreaders. and a
reception beam generator 145 for generating a reception beam using the weight
vector calculated by the signal processor 144.
A lirst multipath signal output from the lirst multipath searcher 113 is
input to the lirst despreader 141. The lirst despreader 141 despreads the first
multipath signal output from the lirst multipath searcher 113 with a predetermined
despreading code, and outputs the despread multipath signal to the signal
processor 144 and the reception beam generator 145. Here, the despreading
process is called "temporal processing." Similarly, a first multipath signal output
from the second multipath searcher 123 is input to the second despreader 142.
The second despreader 142 despreads the lirst multipath signal output from the
second multipath searcher 123 with a predetermined despreading code, and
outputs the despread multipath signal to the signal processor 144 and the
reception beam generator 145. Similarly, a first multipath signal output from the
Nth multipath searcher 133 is input to the Nth despreader 143. The Nth despreader
143 despreads the first multipath signal output from the Nth multipath searcher
133 with a predetermined despreading code, and outputs the despread multipath
signal to the signal processor 144 and the reception beam generator 145.
The signal processor 144 receives the signals output from the first
despreader 141 to the Nth despreader 143. and calculates a reception beam weight
set W_k for generating a reception beam. A set of first multipath signals output

from the first multipath searcher 113 to the Nth multipath searcher 133 will be
defined as " X R." The first multipath signal set X R represents a set of first
multipath signals received via the first reception antenna 111 to the Nth reception
antenna 131 at a kth point, and the first multipath signals constituting the first
multipath signal set X R are all vector signals. The reception beam weight set
WR represents a set of reception beam weights to be applied to the first multipath
signals received via the first reception antenna 111 to the Nth reception antenna
131 at the kth point, and the reception beam weights constituting the weight set
WR are all vector signals.
A set of signals determined by despreading all of the first multipath
signals in the first multipath signal set X R will be defined as yR . The despread
signal set YR. of the first multipath signals represents a set of signals determined
by despreading the first multipath signals received via the first reception antenna
111 to the Nth reception antenna 131 at the kth point, and the despread signals
constituting despread signal set yR of the first multipath signals are all vector
signals. Herein, for the convenience of explanation, the term "set" will be omitted,
and the underlined parameters represent sets of corresponding elements.
Each of the first despreaders 141 to the Nth despreaders 143 despreads the
first multipath signal XR with a predetermined despreading code, such that the
reception power of a desired reception signal is greater than the reception power
of an interference signal by a process gain. The despreading code is identical to
the spreading code used in the transmitters of the MSs.
As described above, the despread signal y of the first multipath signal
XR is input to the signal processor 144. The signal processor 144 calculates a
reception beam weight WR with the despread signal Y of the first multipath
signal XR and outputs the reception beam weight IT. to the reception beam
generator 145. As a result, the signal processor 144 calculates the reception beam
weight WR including a total of N weight vectors applied to the first multipath
signal XR. output from the first reception antenna 111 to the Nth reception
antenna 131. with the despread signals yR of a total of N first multipath signals

oulpul from the first reception antenna 111 to the Nth reception antenna 131. The
reception beam generator 145 recehes the despread signals yR of a total of the N
first multipath signals XR and a total of the N reception beam weight vectors
WR. The reception beam generator 145 generates a reception beam with a total of
the N reception beam weight vectors W. calculates a scalar product of the
despread signal yR of the first multipath signal X and the reception beam
weight WR corresponding to the reception beam, and outputs the result as an
output zk of the first finger 140-1. The output zk of the first linger 140-1 can be
expressed as shown in Equation (1).
Zk - Wk H yk (1)
In Equation (1). H denotes a Hermitian operator, i.e.. a conjugate-
transpose. A set zk. of output signals zk from I. fingers in the BS receiver is
finally input to the multipath combiner 150.
Although only the first finger 140-1 has been described, the other fingers,
the second linger 140-2 to the Lth finger 140-1., are to the same as the first finger
140-1 in operation.
The multipath combiner 150 combines the signals oulpul from the first
finger 140-1 to the Lth linger 140-1., and outputs the combined signal to the
deinterleaver 160. The deinterleaver 160 deinterleaves the signal output from the
multipath combiner 150 in a deinterleaving method corresponding to the
interleaving method used in the transmitter, and outputs the deinterleaved signal
to the decoder 170. The decoder 170 decodes the signal output from the
deinterleaver 160 in a decoding method corresponding to the encoding method
used in the transmitter, and outputs the decoded signal as final reception data.
The signal processor 144 calculates a reception beam weight WR such
that a Mean Square Error (MSH) of a signal received from a MS transmitter,
desired to be received by a predetermined algorithm, becomes minimized. The
reception beam generator 145 generates a reception beam using the reception

beam weight WR, generated by the signal processor 144. The process of
generating a reception beam such that the MSI:, is minimized is called "spatial
processing." When the Rx-AAA scheme is used in a CDMA mobile
communication system, temporal processing and spatial processing are
simultaneously performed. The operation of simultaneously performing temporal
processing and spatial processing is called "spatial-temporal processing.*'
The signal processor 144 receives multipath signals despread for each
linger in the above-described manner, and calculates a reception beam weight
capable of maximizing a gain of the Rx-AAA scheme according to a
predetermined algorithm. The signal processor 144 minimizes the MSE.
Currently, a great deal of research is being conducted on a reception beam
weight calculation algorithm for adaplively minimizing the MSL:1. However, the
reception beam weight calculation algorithm for adaptively minimizing the MSE
is an algorithm for reducing errors on the basis of a reference signal, and this
algorithm supports a Constant Modulus (CM) scheme and a Decision-Directed
(DD) scheme as a blind scheme, when there is no reference signal.
further, the algorithm for reducing errors on the basis of a reference
signal has trouble converging into a minimum MSI value desired by the system
in an environment where a channel such as a fast fading channel suffers from a
rapid change, or an environment where a high-order modulation scheme such as
16-ary quadrature amplitude modulation (16QAM) is used. liven though it
converges into a particular MSE value, the minimum MSE value is set to a
relatively large value. When the minimum MSE value is set to a relatively large
value, a gain that occurs from using the Rx-AAA scheme is reduced. Therefore,
this algorithm is not suitable for a high-speed data communication system.
SUMMARY OF THE INVENTION
It is, therefore, an object of the present invention to provide an apparatus
and a method for receiving data using an Adaptive Antenna Array scheme in a
mobile communication system.

It is another object of the present invention to provide an apparatus and a
method for receiving data using an adaptive reception beam weight generation
scheme in a mobile communication system using an Adaptive Antenna Array
scheme.
It is further another object of the present invention to provide an apparatus
and a method for generating a reception beam having a minimum error value in a
mobile communication system using an Adaptive Antenna Array scheme.
In accordance with a first aspect of the present invention, there is
provided a method of generating a reception beam weight for generating a
reception beam from a reception signal, the method comprisingdelermining a first
error value by using a first scheme at a timing point, and a second error value by
using a second scheme different from the first scheme at the liming point,
determining a first scheme application weight according to a difference between
the first error value and the second error value, and a second scheme application
weight according to the difference between the first error value and the second
error value, generating a third error value using a scheme that combines the first
scheme to which the first scheme application weight is used and the second
scheme to which the second scheme application weight is used, determining a
reception beam weight using the reception signal, the third error value, and an
output signal generated by using the reception beam to the reception signal,
wherein the reception beam weight is used for generating the reception beam.
In accordance with a second aspect of the present invention, there is
provided an apparatus of generating a reception beam weight for generating a
reception beam from a reception signal, the apparatus comprising an error value
combiner of determining a first error value by using a first scheme at a liming
point, and a second error value by using a second scheme different from the first
scheme at the timing point, determining a first scheme application weight
according to a difference between the first error value and the second error value,
and a second scheme application weight according to the difference between the
first error value and the second error value: generating a third error value using a
scheme that combines the first scheme to which the first scheme application
weight is used and the second scheme to which the second scheme application

weight is used, a weight calculator of determining a reception beam weight using
the reception signal, the third error value, and an output signal generated by using
the reception beam to the reception signal, wherein the reception beam weight is
used for generating the reception beam.
BRIEF DESCRIPTION OF THE ACCOMPANYING DRAWINGS
The above and other objects, features, and advantages of the present
invention will become more apparent from the following detailed description
when taken in conjunction with the accompanying drawings in which:
FIG. 1 is a block diagram illustrating a base station receiver in a
conventional CDMA mobile communication system:
FIG. 2 is a graph illustrating a characteristic of sigmoid function used in
an embodiment of the present invention:
FIG. 3 is a block diagram illustrating a base station receiver according to
an embodiment of the present invention:
FIG. 4 is a flowchart illustrating a signal reception procedure by a base
station receiver according to an embodiment of the present invention:
FIG. 5 is a diagram illustrating Constant Modulus (CM) schemes in an
OFDM mobile communication system:
FIG. 6 is a diagram schematically illustrating Decision Directed (DD)
schemes in an OFDM mobile communication system using Binary Phase Shift
Keying (BPSK):
FIG. 7 is a graph illustrating a characteristic curve for a general reception
beam weight generation schemes and a reception beam weight generation scheme
according to an embodiment of the present invention:
FIG 8 is a graph illustrating a characteristic curve according to the
number of reception antennas of a base station receiver for an reception beam
weight generation scheme according to an embodiment of the present invention:
and
FIG. 9 is a block diagram illustrating a structure of an OFDM mobile
communication system according to an embodiment of the present invention.
DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT

Several preferred embodiments of the present invention will now be
described in detail herein below with re Terence to the annexed drawings. In the
drawings, the same or similar elements are denoted by the same reference
numerals even though they are depicted in different drawings. In the following
description, a detailed description of known functions and configurations
incorporated herein has been omitted for conciseness.
Before a description of the present invention is given, a model of a
reception signal received at a receiver of a base station (BS) will be considered. It
is assumed that a receiver of the BS includes a receive-antenna array having a
plurality of reception antennas (Rx ANTs), and the receive-antenna array is
generally mounted only in the receiver of the BS considering its cost and size, and
is not mounted in a receiver of a mobile station (MS). That is. it is assumed that
the receiver of the MS includes only one reception antenna.
Additionally, although the present invention is applicable to all of mobile
communication systems using Frequency Division Multiple Access (FDMA).
Time Division Multiple Access (TDMA). Code Division Multiple Access
(CDMA), and Orthogonal Frequency Division Multiplexing (OFDM), the present
invention will be described with reference to an OFDM mobile communication
system.
A signal transmitted from a transmitter of an mth MS existing in a cell
serviced by the BS is expressed as shown in Equation (2).

In Kquation (2), sm(t) denotes a transmission signal of an m11 MS. pm
denotes transmission power of the mth MS. bm(t) denotes a user information bit
sequence of the mth MS. and em(t) denotes a user spreading code sequence of the
mth MS. having a chip period of Tc.
The transmission signal transmitted from the MS transmitter is received at
a receiver of the BS over a multipath vector channel. It is assumed that channel
parameters of the multipath vector channel are relatively and continuously

changed, compared with the bit period Tb. Therefore, it is assumed that the
channel parameters of the multipath vector channel are constant for certain bit
periods.
A complex base band reception signal for a first multipath of an mth MS.
received at a receiver of the BS. is expressed by liquation (3). It should be noted
that the reception signal of liquation (3) represents a base band signal determined
by down-converting a radio frequency (RF) signal received at the BS receiver.

In Equation (3). denotes a set of complex base band reception
signals received through a first multipath of the mth MS. αm1 denotes a fading
attenuation applied to the first multipath of the mth MS. Φm1 denotes a phase
transition applied to the first multipath of the mth MS. τm1 denotes a lime delay
applied to the first multipath of the mth MS. and denotes a set of array
responses (ARs) applied to the first multipath of the mth MS. Because the BS
receiver includes a plurality of antennas, for example. N reception antennas, a
signal transmitted by the mth MS is received at the BS receiver via the N reception
antennas. Therefore, the number of signals received via the first multipath is N.
and N complex base band reception signals received via the first multipath of the
mth MS constitute a set of the reception signals. Herein, for the convenience of
explanation, the term "set" will be omitted, and the underlined parameters
represent sets of corresponding elements.
When a current linear antenna array is used, the array response am1 is
defined as shown in Equations (4).

In Equation (4). 'd' denotes a distance between separated reception
antennas. λ denotes a wavelength at a frequency band in use. N denotes the

number of the reception antennas, and 0m1 denotes direction-of-arrival (DOA)
applied to the first multipath of the mth MS.
If it is assumed that the number of MSs existing in a cell serviced by the
BS is M and there are L multiple paths for each of the M MSs. a reception signal
received at the BS becomes the sum of transmission signals transmitted from the
M MSs and additive white noise (AWN), as represented in liquation (5).

In Equation (5). n(l) denotes the additive while noise added to the
transmission signals transmitted from the M MSs.
It is assumed that a signal the BS desires to receive in the reception signal
of Hquation (5) is x11, x11 represents a signal a first MS has transmitted via a
first multipath. Because it is assumed thai a signal the BS desires to receive is x11.
all signals except the signal x11 are regarded as interference signals and noise.
Therefore. liquation (5) can be rewritten as shown in liquation (6).

In liquation (6), i(l) denotes an interference signal, which is defined in
liquation (7). below.

The first term of Equation (7) is a transmission signal of a MS
that the BS desires to receive, but represents the inter-path interference (IPI) by
other multiple paths lhat the BS does not desire to receive. The second term

(7) represents the multiple access interference (MAI) by
Further, the x(l) is desprcad with a desprcading code C1(1-τ11) previously
set in a first finger (1-1) for a corresponding multipath in a corresponding channel
card of the BS receiver, i.e.. a channel card (m-1) assigned to the first MS. and
the desprcad signal y(l) is defined in Fquation (8) below. The despreading code
C1(1-τ11) is identical to the despreading code c,(t-Tn) used in a BS transmitter
during signal transmission. The BS includes a plurality of receivers as described
in conjunction with FIG. 1. Hach of the receivers is called a "channel card." and
one channel card is assigned to one MS. As described in connection with FIG. 1.
the channel card includes as many lingers as the number of multiple paths, and
the fingers are mapped to corresponding multipath signals on a one-to-one basis.

When the signal y(l) is generated by despreading the pre-despread
signal x(l) with the despreading code C1(1-τ11), the power of a signal component
the BS receiver desires to receive from among the reception signals is amplified
by a gain G according to a characteristic of a despreader. It is noted that although
the power of a signal component the BS receiver desires to receive is amplified
by a process gain G. the power of the signal components the BS receiver does not
desire to receive is not changed at all. Therefore, a correlation matrix between a
reception signal before despreading and a reception signal after despreading can
be calculated.
In order to calculate the correlation matrix between a reception signal
before despreading and a reception signal after despreading. the reception signal
x(l) before despreading is sampled at a kth point, which is equal to the sampling
point of the reception signal y(l) after desprcading. The signal obtained by

sampling the reception signal x(l) before despreading at the kth point is
represented by Equation (9).

In conclusion, in order to calculate a correlation matrix between a
reception signal x(l) before despreading and a reception signal y(l) after
despreading. it is assumed that the signal of liquation (9) is acquired by sampling
the reception signal x(l) before despreading at the kth point, which is equal to
the sampling point of the reception signal y(l) after despreading. and that the
reception signal x(l) before despreading and the reception signal y(l) after
despreading are stationary.
A description will now be made of a Least Mean Square (LMS) scheme
and a Minimum Mean Square Error (MMSE) scheme herein below.
In the LMS scheme, a set of reception signals belbre despreading.
including complex reception signals received via N reception antennas at a
particular time. i.e.. complex reception signals x1 to xN received via a first
reception antenna to an Nth reception antenna, will be defined as x -
[X1,X2,...,XN]T. Here. "T" is an operator representing a transpose operation. In
addition, a set of reception signals after despreading the complex reception
signals X1, X2.... xN received via the N reception antennas will be defined as
y=[y1,y2,... ,yN]T The reception signal y after despreading is determined by the
sum of a signal component s the BS receiver desires to receive and a signal
component u the BS receiver does not desire to receive, as represented by
liquation (10).

A set of complex reception beam weight values to be multiplied by the
complex reception signals x1, x2,..., xN received via the N reception antennas, i.e..
complex reception beam weights w1 to wN to be multiplied by complex reception

signals x1 to xN received via the first reception antenna to the Nth reception
antenna, will be defined as
An output signal z from lingers in a particular user card. i.e.. a channel
card assigned to a particular MS. is determined by calculating a scalar product of
the reception beam weight w and the reception signal y after despreading. as
represented by Equation (11).

In liquation (11). 'i' denotes the number of reception antennas.
The output signal z can be classified into a signal component wII s the
BS receiver desires to receive, and a signal component wII u the BS receiver
does not desire to receive, using liquation (10) and liquation (11). The LMS
scheme minimizes errors of a known reference signal and a reception signal, and
particularly, minimizes a cost function J(w). as given below in equation (12).

In liquation (12). 'J' denotes a cost function, and a reception beam weight
value w for minimizing the cost function value J must be determined, further, in
Hquation (12). ck denotes a difference, or an error, between a reception signal and
a desired reception signal, and dk denotes the desired signal. In a reception beam
algorithm using a non-blind scheme, a pilot signal is used as the desired signal dk
by way of example. However, the present invention proposes a reception beam
algorithm using a blind scheme, such that a detailed description of the reception
beam algorithm using the non-blind scheme will be omitted.
In liquation (12). the cost function J is a type of a second-order convex
function. Therefore, in order to minimize the cost function J. the cost function J

must be differentiated until its value becomes 0. A differentiated value of the cost
function .) is shown below in Equation (13).

However, it is difficult to acquire an optimal reception beam weight wII
in an actual channel environment in a single process, and because the reception
signal y after despreading is input at each point, a recursive formula of
liquation (14) should be used in order to adaptive!}' or recursively acquire the
optimal reception beam weight wII.

In Hqualion (14), 'k' denotes a kth point. wR denotes a reception beam
weight at the kth point, µ denotes a constant gain, and vR denotes a trace vector
at the kth point. The trace vector v at the kth point represents a vector for
converging a differentiated value of the cost function J to a minimum value, for
example. 0. That is. liquation (14) shows a process of updating a value generated
before or after a constant gain µ from a given reception beam weight wR to be
used at a current point in a direction of the trace vector vR as a reception beam
weight wR-1 to be used at the next point.
The scheme for delecting a desired reception signal d(k). proposed in the
present invention, is called a "blind scheme." Due to the use of the blind scheme,
a received signal should be adaptively converged using a particular estimation
value, and a below scheme is used for the adaptive convergence of the received
signal.
A combination mode blind scheme is used for detecting a desired received
signal d(k). In this case, an error function is expressed as shown in liquation (15).

In Equation (15). arc delected error values by applying
Constant Modulus (CM) scheme and Decision Directed (DD) scheme to the
received signal for an adaptive convergence of the received signal. The
will be described herein below.
In the present invention, as indicated above. are detected by
applying combination of the CM scheme and the DD scheme to the received
signal for an adaptive convergence of the received signal. That is. when the
value is increased, a value is also increased. Consequently, an influence
of the DD scheme is increased to a total error value.
In Equation (15). g(x) is an s shape-function (sigmoid function).
Accordingly, in a region with a large influence of the CM scheme, an influence of
the DD scheme is decreased. In contrast, in a region with a large influence of the
DD scheme, an influence of the CM scheme is decreased.
In Hqualion (15). the error value ek is an error value by combining a value
by applying a weight ak to the ekCM and a value by applying a weight P to
the ekDD. Herein, the weight ak is a weight applied to the CM scheme as -CM
scheme application weight, and the weight βk is a weight applied to the DD
scheme as "DD scheme application weight'. Therefore, the error value ek is an
error value detected by setting adaptively the weight ak and the weight P
according to whether or not the error value of the received signal was converged.

Additionally, the characteristic of sigmoid function g(x) will be described
herein below.
The FIG. 2 is a graph illustrating the characteristic of sigmoid function
used in an embodiment of the present invention. Referring to FIG. 2. the
characteristic of sigmoid function is changed according to value 'a'. When the
value a increases, a shape of sigmoid function is closed to an 's' shape. When the
value a is equal to 1 (a _ 1). the shape of sigmoid function is closed to a
"straight line" shape. That is, when the CM scheme application weight ak is
increased, the DD scheme application weight βk is decreased. However, when
the CM scheme application weight αk is decreased, the DD scheme application
weight βk is increased.
If the error value ek calculated using the CM scheme is greater than the
error value ek calculated using the DD scheme, the error value ek will be
calculated by combining a weighted CM scheme and a more weighted DD
scheme, compared with the weighted CM scheme.
Herein, the weighted CM scheme is used to apply the CM scheme
application weight αk to the CM scheme. The weighted DD scheme is used to
apply the DD scheme application w eight βk to the DD scheme. The "more
weighted DD scheme as compared with the weighted CM scheme" refers to the
DD scheme application weight βk being greater than the CM scheme application
weight αk. In the same manner, the "more weighted CM scheme as compared
with the weighted DD scheme refers" to the CM scheme application weight a
being greater than the DD scheme application weight βk.
Additionally, if the error value ek calculated by using the CM scheme is
equal or smaller than the error value ek calculated by using the DD scheme, the
error value ek will be calculated by combining a weighted DD scheme and a more
weighted CM scheme, compared with the weighted DI) scheme.
A constant modulus (CM) scheme, which is used for adaptive

convergence of the received signal, is generally used in a blind equalizer and also
used for a generation algorithm. When the CM scheme proposed by Godard is
used, the cost function J is expressed as shown in Equation (16) below.

In liquation (16). 'p' denotes a particular positive integer, and Rp denotes
a Godard modulus. The Godard modulus Rp is defined as shown in liquation (17).

Because the current OFDM mobile communication system generally uses
a high-order modulation scheme, which is higher in order than quadrature phase
shift keying (QPSK) modulation, the cost function .1 is separated into a real part
and an imaginary part as shown in liquation (18). The cost function .1 is separated
into a real part and an imaginary part because transmission reception signals in
the high-order modulation scheme have a real part and an imaginary part.

It is assumed herein that the present invention uses the FMS scheme and
the MMSF. scheme, and p-2. Therefore. d(k)=R2.R+jR2.1. In addition, it is assumed
that a cost function value J at an initial point, i.e.. a k=0 point, is 0 (J=0).
FIG. 5 is a diagram illustrating a CM scheme in an OFDM mobile
communication system. Referring to FIG. 5. a CM scheme for p=2. d(k)-
R2.R-jR2.1, and J-0 at a point with k-0. That is, if a value R2 is determined by

liquation (18). a circle is generated on a coordinate surface. Then, a received
signal is determined as a point where an extension line drawn from the origin
meets the circle. In FIG. 5. received zk is projected as a circle.
Above, the convergence step has been described. Herein below, a
stabilizing step for acquiring the d(k) will be described.
If MSE is converged into a predetermined value through the convergence
step, a change occurs from the convergence step to the stabilization step where
calculation of Equation 19 is performed. A process where a change occurs from
the convergence step to the stabilization step as the MSF. is converged into a
predetermined value will be described later on.

In the stabilization step, like in the convergence step, a real part and an
imaginary part are separately calculated. In liquation (19). Pr denotes a received
signal is projected as a signal most approximating the desired reception signal
d(k) by a DD scheme. The DD scheme is a scheme for reflecting the d(k) as a
decision value most approximating the received signal.
FIG. 6 is a diagram illustrating a DD scheme in an OFDM mobile
communication system using Binary Phase Shift Keying (BPSK). Referring to
FIG. 6, because it is assumed that the 01 DM mobile communication system uses
BPSK. if a reception signal is (1.2. -0.2) in an I-Q domain, the desired reception
signal d(k) is projected as the largest approximate value of 1 after calculating a
distance from - 1 and -1.
FIG. 3 is a block diagram illustrating a BS receiver according to a first
embodiment of the present invention. While describing FIG. 3. it should be noted
that a BS receiver according to the First embodiment of the present invention is
similar in structure to the BS receiver described in connection with FIG. 1. but
different in a method for determining a reception beam weight by a signal
processor. For simplicity, only the elements directly related to the present

invention in the BS receiver will be described with reference to FIG. 3. Further,
the first embodiment of the present invention corresponds to an embodiment
where the LMS scheme is used.
Referring to FIG. 3. when a reception signal x, at a timing point k is
received, a despreader 310 despreads the reception signal xk, using a
predetermined despreading code, and outputs the despread reception signal y
to a signal processor 330 and a reception beam generator 320. The signal
processor 330 includes a weight calculator 331. a memory 333. and an error value
combiner 335. For simplicity. FIG. 3 will be described with reference to only the
first finger 140-1 in the BS receiver of FIG. 1. Therefore, the desprcader 310 of
FIG. 3 is substantially identical in operation to the N despreaders of the first
despreader 141 to the Nth desprcader 143 in the first finger 140-1.
The error value combiner 335 inputs the despread reception signal y .
and combines an error value ek by using the CM scheme and the DD scheme. The
weight calculator 331 in the signal processor 330 calculates a reception beam
weight wR by receiving the combined error value ek . the despread reception
signal yk a predetermined constant gain µ. and an initial reception beam weight
w0. and outputs the calculated reception beam weight to the memory 333. The
memory 333 buffers the reception beam weight wR calculated by the weight
calculator 331, and the weight calculator 331 uses the reception beam weight wR
stored in the memory 333 when updating the reception beam weight wR. That is.
the weight calculator 331 updates a reception beam weight wR. at the next
timing point k+1 using the reception beam weight wR calculated at the timing
point k.
FIG. 4 is a flowchart illustrating a signal reception procedure by a BS
receiver according to an embodiment of the present invention. Referring to FIG 4.
in step 411. a BS receiver establishes an initial reception beam weight it, . and a
constant gain µ. In step 413. the BS receiver determines if the communication has
ended. If it is determined that the communication has ended, the BS receiver ends
the ongoing procedure.

If it is determined in step 413 that the eommunieation has not ended, the
BS receiver proceeds to step 415. In step 415, the BS receiver receives a despread
signal yk for the reception signal x. In step 417. the BS receiver calculates a
set zk of signals zk output from respective fingers of the BS receiver using the
despread signal yk and a reception beam weight The zR
represents a set of finger output signals generated using a reception beam
generated using the reception beam weight wR.
In step 419. the BS receiver calculates an error value ek to decrease an
error between the reception signal xk and a desired reception signal
. In step 421. the BS receiver calculates a
differentiated value of a cost function using the despread signal y. and the error
function ek (VJ(wR) = -2ekyk). In step 423. the BS receiver calculates a reception
beam coefficient, or a reception beam weight
In step 425. the BS receiver maintains the calculated reception beam
weight wR. In step 427. the BS receiver delays by a predetermined unit lime. The
predetermined unit time is delayed in order to use a value determined at a kth snap
for a (k-1)th snap. i.e.. to take a slate transition delay into consideration. In step
429. the BS receiver increases the k by 1. i.e.. transitions from the current timing
point k to the next timing point k+1. and then returns to step 413.
FIG. 7 is a graph illustrating a characteristic curve for a general reception
beam weight generation schemes and an reception beam weight generation
scheme according to an embodiment of the present invention. Referring to FIG. 7.
it is noted that an MSF value (y-axis) compared to a number of iterations (x-axis)
for the reception beam weight generation scheme according to the present
invention 703 is converged into a lower value, compared with an MSF value
against the number of iterations for the conventional reception beam weight
generation scheme 701, e.g.. a CM scheme. That the MSF value is converged into
a less value means that a reception beam can be correctly generated, making it

possible to correctly receive only a desired reception signal.
FIG. 8 is a graph illustrating a characteristic curve according to the
number of reception antennas of a BS receiver tor an adaptive reception beam
weight generation scheme according to the embodiment of the present invention.
Referring to FIG. 8. there is illustrated a radiation pattern for a BS receiver having
6 reception antennas and a BS receiver having 10 reception antennas. For
example, if it is assumed that a particular BS is located at 57°. it is noted that
compared with the BS receiver having 6 reception antennas, the BS receiver
having 10 reception antennas has a normalized antenna gain of about 0.2. and can
more correctly generate a reception beam. As a result, in terms of capacity of an
OFDM mobile communication system, an increase in the number of the reception
antennas causes an increase in the amplitude of the reception signals enabling a
correct communication, thereby contributing to an increase in system capacity.
FIG. 9 is a block diagram illustrating a structure of an OFDM mobile
communication system according to an embodiment of the present invention.
Referring to FIG. 9. the OFDM communication system includes a transmitter, i.e..
an MS transmitter 900. and a receiver, i.e.. a BS receiver 950. The MS transmitter
900 includes a symbol mapper 911, a serial-to-parallel (or S P) converter 913. a
pilot pattern inserter 915. an inverse fast Fourier transform (IFFT) unit 917. a
parallel-to-serial (or P/S) converter 919. a guard interval inserter 921. a digital-to-
analog (D/A) converter 923. and a radio frequency (RF) processor 925.
When there are information data bits to be transmitted, the information
data bits are input to the symbol mapper 911. The symbol mapper 911 modulates
the input information data bits in a predetermined modulation scheme for symbol
mapping, and outputs the symbol-mapped data bits to the serial-to-parallel
converter 913. Here, quadrature phase shift keying (QPSK) or 16-ary quadrature
amplitude modulation (I6QAM) can be used as the modulation scheme. The
serial-to-parallel converter 913 parallel-converts serial modulation symbols
output from the symbol mapper 911. and outputs the parallel-converted
modulation symbols to the pilot pattern inserter 915. The pilot pattern inserter 915
inserts pilot patterns in the parallel-converted modulation symbols output from
the serial-to-parallel converter 913. and then outputs the pilot pattern-inserted

modulation symbols to the IFFT unit 917.
The IFFT unit 917 performs N-point IFFT on the signals output from the
pilot pattern inserter 915. and outputs the resultant signals to the parallel-to-serial
converter 919. The parallel-to-serial converter 919 serial-converts the signals
output form the IFFT unit 917. and outputs the serial-convened signals to the
guard interval inserter 921. The guard interval inserter 921 receives the signal
output from the parallel-to-serial converter 919, inserts a guard interval therein,
and outputs the guard interval-inserted signal to the digilal-to-analog converter
923.
The guard interval is inserted to remove interference between a previous
OFDM symbol transmitted at a previous OFDM symbol time and a current
OFDM symbol to be transmitted at a current OFDM symbol time in an 01 DM
communication system. For the guard interval, a cyclic prefix method or a cyclic
postfix method is used. In the cyclic prefix method, a predetermined number of
last samples of an OFDM symbol in a lime domain are copied and inserted into a
valid OFDM symbol. In the cyclic postfix method, a predetermined number of
first samples of an OFDM symbol in a lime domain arc copied and inserted into a
valid OFDM symbol.
The digilal-to-analog converter 923 analog-converts the signal output
from the guard interval inserter 921. and outputs the analog-converted signal to
the RF processor 925. The RF processor 925. including a tiller and a front-end
unit (not shown). RF-proccsses the signal output from the digital-to-analog
converter 923 such that the signal can be transmitted via an antenna.
The BS receiver 950 includes an RF processor 951. an analog-to-digital
(or A/D) converter 953. a reception beam generator 955. a signal processor 957. a
guard interval remover 959. a serial-to-parallel (or SP) converter 961. a fast
Fourier transform (FFT) unit 963. an equalizer 965. a pilot symbol extractor 967.
a synchronization & channel estimation unit 969. a parallel-to-serial (or P S)
converter 971. and a symbol demapper 973.
The signals transmitted by the MS transmitter 900 are received via

reception antennas of the BS receiver 950. The received signals experience a
multipath channel and have a noise component. The signals received via the
reception antennas are input to the RF processor 951. which down-converts the
signals received via the reception antennas into an intermediate frequency (IF)
signal, and outputs the IF signal to the analog-to-digital converter 953. The
analog-to-digital converter 953 digital-converts an analog signal output from the
RF processor 951. and outputs the digital-converted signal to the reception beam
generator 955 and the signal processor 957. Operations of the reception beam
generator 955 and the signal processor 957 have been described above with
reference to the first and second embodiments of the present invention. Therefore,
a detailed description thereof will not be given again.
The signal output from the reception beam generator 955 is input to the
guard interval remover 959. The guard interval remover 959 removes a guard
interval from the signal output from the reception beam generator 955. and
outputs the resultant signal to the serial-to-parallel converter 961. The serial-to-
parallel converter 961 parallel-converts the serial signal output from the guard
interval remover 959. and outputs the resultant signal to the IF 1 unit 963. The
FFT unit 963 performs N-point FFT on the signal output from the serial-to-
parallel converter 961. and outputs the resultant signal to the equalizer 965 and
the pilot symbol extractor 967.
The equalizer 965 performs channel equalization on the signal output
from the FFT unit 963. and outputs a resultant signal to the parallel-to-serial
converter 971. The parallel-to-serial converter 971 serial-converts the parallel
signal output from the equalizer 965. and outputs a resultant signal to the symbol
demapper 973. The symbol demapper 973 demodulates the signal output from the
parallel-to-serial converter 971 using a demodulation scheme corresponding to
the modulation scheme used in the MS transmitter 900. and outputs a resultant
signal as received information data bits.
The signal output from the FFT unit 963 is input to the pilot symbol
extractor 967. and the pilot symbol extractor 967 extracts pilot symbols from the
signal output from the FFT unit 963. and outputs the extracted pilot symbols to
the synchronization & channel estimation unit 969. The synchronization &

channel estimation unit 969 synchronizes and channel estimates the pilot symbols
output from the pilot symbol extractor 967, and outputs the result to the equalizer
965.
As is understood from the foregoing description, the mobile
communication system according to the present invention generates a weight
using an adaptive reception beam weight generation scheme combined a CM
scheme and a DD scheme, thereby making it possible to rapidly generate a
reception beam weight with a minimum error value. Therefore, it is possible to
generate a correct reception beam, and the correct reception of a reception beam
enables a receiver to correctly receive only a desired signal, thereby improving
system performance.
While the present invention has been shown and described with reference
to certain preferred embodiments thereof, it will be understood by those skilled in
the art that various changes in form and details may be made therein without
departing from the spirit and scope of the invention as defined by the appended
claims.

WE CLAIM:
1. A method for generating a reception beam weight for generating a
reception beam from a reception signal, the method comprising the steps of:
determining a first error value by using a Constant Modulus (CM)
scheme at a timing point, and a second error value by using a Decision-
Directed (DD) scheme different from the CM scheme at the timing point
(419);
determining a CM scheme application weight and a DD scheme
application weight differently by comparing with the first error value and the
second error value;
generating a third error value using a scheme that combines the CM
scheme to which the CM scheme application weight is applied and the DD
scheme to which the DD scheme application weight is applied (421);
determining a reception beam weight using the reception signal, the third
error value, and an output signal generated by applying the reception beam
to the reception signal, wherein the reception beam weight is used for
generating the reception beam (423).
2. The method as claimed in claim 1, wherein the CM scheme application
weight and the DD application scheme application weight are determined
using sigmoid function.

3. The method as claimed in claim 1, wherein the DD scheme application
weight is determined as a value greater than the CM scheme application
weight, if the first error value is greater than the second error value.
4. The method as claimed in claim 1, wherein the CM scheme application
weight is determined as a value greater than the DD scheme application
weight, if the first error value is not greater than the second error value.
5. The method as claimed in claim 1, wherein each of the first and the
second the error values is a value representative of a difference between a
desired reception signal and the output signal.
6. The method as claimed in claim 5, wherein each of the first and the
second the error values is a mean square error (MSE) value.
7. An apparatus for generating a reception beam weight for generating a
reception beam from a reception signal, the apparatus comprising :
an error value combiner (335) of determining a first error value by using a
Constant Modulus (CM) scheme at a timing point, and a second error value
by using a Decision Directed (DD) scheme different from the CM scheme at

the timing point, determining a CM scheme application weight and a DD
scheme application weight differently by comparing with the first error
value and the second error value; generating a third error value using a
scheme that combines the CM scheme application weight is applied and the
DD scheme to which the DD scheme application weight is applied ;
a weight calculator (331) of determining a reception beam weight using
the reception signal, the third error value, and an output signal generated by
applying the reception beam to the reception signal, wherein the reception
beam weight is used for generating the reception beam.
8. The apparatus as claimed in claim 7, wherein the error value combiner
comprises means for determining the CM scheme application weight and
the DD application scheme application weight using sigmoid function.
9. The apparatus as claimed in claim 7, wherein the error value combiner
comprises means for determining the DD scheme application weight greater
than the CM scheme application weight, if the first error value is greater
than the second error value.

10. The apparatus as claimed in claim 7, wherein the error value combiner
comprises means for determining the CM scheme application weight greater
than the DD scheme application weight, if the first error value is not greater
than the second error value.
11. The apparatus as claimed in claim 7, wherein each of the first and the
second the error values is a value representative of a difference between a
desired reception signal and the output signal.
12. The apparatus as claimed in claim 1 1, wherein each of the first and the
second the error values is a mean square error (MSE) value.

A method for generating a reception beam weight for generating a reception
beam from a reception signal, the method comprising the steps of
determining a first error value by using a Constant Modulus (CM) scheme at
a timing point, and a second error value by using a Decision-Directed (DD)
scheme different from the CM scheme at the timing point (419); determining
a CM scheme application weight and a DD scheme application weight
differently by comparing with the first error value and the second error
value; generating a third error value using a scheme that combines the CM
scheme to which the CM scheme application weight is applied and the DD
scheme to which the DD scheme application weight is applied (421);
determining a reception beam weight using the reception signal, the third
error value, and an output signal generated by applying the reception beam
to the reception signal, wherein the reception beam weight is used for
generating the reception beam (423).

Documents:

2430-KOLNP-2005-FORM-27.pdf

2430-kolnp-2005-granted-abstract.pdf

2430-kolnp-2005-granted-claims.pdf

2430-kolnp-2005-granted-correspondence.pdf

2430-kolnp-2005-granted-description (complete).pdf

2430-kolnp-2005-granted-drawings.pdf

2430-kolnp-2005-granted-examination report.pdf

2430-kolnp-2005-granted-form 1.pdf

2430-kolnp-2005-granted-form 18.pdf

2430-kolnp-2005-granted-form 3.pdf

2430-kolnp-2005-granted-form 5.pdf

2430-kolnp-2005-granted-gpa.pdf

2430-kolnp-2005-granted-reply to examination report.pdf

2430-kolnp-2005-granted-specification.pdf

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Patent Number

231423

Indian Patent Application Number

2430/KOLNP/2005

PG Journal Number

10/2009

Publication Date

06-Mar-2009

Grant Date

04-Mar-2009

Date of Filing

30-Nov-2005

Name of Patentee

SAMSUNG ELECTRONICS CO., LTD.

Applicant Address

416, MAETAN-DONG, YEONGTON-GU, SUWON-SI, GYEONGGI-DO

Inventors:

#	Inventor's Name	Inventor's Address
1	CHANG-HO SUH	14-15, DAEBANG-DONG, DONGJAK-GU, SEOUL
2	CHAN-BYONG CHAE	#104-1701, BYUCKSAN APT JEGI 2-DONG, DONGDAEMUN-GU SEOUL
3	KATZ, MARCOS DANIEL	#621-906, DONGBO APT, YEONGTON-DONG, PALDAL-GU, SUWON-SI, GYEONGGI-DO
4	SEOK-HYUN YOON	# 104-602, HYUNDAI APT, IMUN 3 DONG, DONGDEAMUN-GU SEOUL
5	BYOUNG-YUN KIM	# 201,983-4, YEONGTON-DONG, PALDAL-GU, SUWON-SI, GYEONGGI-DO

PCT International Classification Number

H04B 7/02

PCT International Application Number

PCT/KR2004/001776

PCT International Filing date

2004-07-16

PCT Conventions:

#	PCT Application Number	Date of Convention	Priority Country
1	10/2003-0048898	2003-07-16	Republic of Korea