Title of Invention  METHOD FOR DIRECT CONTROL OF ACTIVE AND REACTIVE POWER FROM THE ROTOR SIDE FOR A GRIDCONNECTED DOUBLYFED SLIPRING INDUCTION MACHINE WITHOUT POSITION ENCODER 

Abstract  A method of direct power control for doublyfed wound rotor induction machine is described. The stator active and reactive powers are controlled within hysteresis bands by adopting a switching algorithm on the rotor side. The control algorithm uses only stator quantities for active and reactive power measurement and is inherently position sensorless. It is computational/y simple and does not involve any machine parameter. A novel method for sector identification of the rotor flux based on the direction of change of reactive power is developed. The algorithm can start 'onthefly' and runs stably even at zero rotor frequency. Fast and decoupled dynamic response of the stator active and reactive powers is achieved. The direct power control method can be used for sensor/ess control of wound rotor induction machines in high power drives and VSCF applications like wind power generation. 
Full Text  Field of Technology This invention relates to the field of power electronics. More particularly this invention relates to control of electrical machines. Precisely this invention relates to a method and system for direct control of active and reactive power from rotor side for a grid connected doublyfed slip ring induction machine without position encoder. Control of doublyfed woundrotor induction machines fed from the rotor side is wellknown and can be effectively used for the control of active and reactive power in motoring or generating applications operating over a limited speedrange. Of late there is an increased attention towards doublyfed induction generators using slipring induction machines for applications like wind electric generation; the primary advantage being reduced rating and cost of converters when they are used on the rotor side. The schematic of such a system is shown in fig.1. The sector is directly connected to the constant frequency grid and the rotor is fed by a variable frequency converter, which can inject slip frequency currents at appropriate phase and magnitude. The rotor side converter can control power in both directions and is interfaced back to the grid by another PWM converter. Such an arrangement provides maximum flexibility in terms of control of active and reactive power. In the existing systems the active power generated and the reactive power drawn from the grid are controlled independently using fieldoriented control technique so that it is possible to operate the system at unity power factor irrespective of the active power generation. The stator flux, rotating in space at the synchronous speed, is taken as the field variable. The rotor currents are first transformed to the field/synchronous reference frame where they are controlled using explicit current loops. The output of the current loops are the reference voltages for the rotor side converter which are then transformed back from the field reference frame to the rotor reference frame. In order to perform the forward and reverse transformations the information of the rotor position with respect to the stator is necessary. The rotor position information in conventional systems is obtained through an incremental or absolute position encoder which is fitted to the rotor shaft. The introduction of this encoder, however, brings down the system reliability. This is particularly disadvantageous in windenergy application where the physical separation between the power electronic equipments (which are at the ground level) and the generator (which is coupled to the turbine) is large. The present trend is therefore, to evolve methods of estimation of rotor position and thereby dispense with the position encoder. Several position sensorless methods have been proposed by different authors in the existing literature. One of the frequently referred methods uses dynamic tracking of the torque angle. Here the rotor voltage is integrated to obtain the rotor flux. The torque angle is then computed from the rotor currents and the rotor flux. This computed angle is forced to track the reference torque angle by using voltagecontrolled oscillators and an integrator. The other methods are based mostly on transformations between the stator and rotor coordinates and can be used with varying degrees of accuracy. Limitations In field oriented control technique, the transient response of active and reactive power is dependent on the degree of decoupling between the field/direct axis and the quadrature axis. This in turn depends on the accuracy of computation of the stator flux magnetising current and accuracy of rotor position information. The former involves machine parameters like stator leakage factor and is prone to some amount of error. Correct alignment of the position encoder is difficult in any doubly fed machine. Here one has to ensure that the zero/reference position of the encoder corresponds to a condition when both the 'a* phase axes of the rotor and stator are in perfect alignment. In order to eliminate the difficulties involving position encoders the sensorless methods as discussed in the previous section are employed. However, since all these methods use machine parameters their accuracy is limited. In the method using rotor voltage integration operation at or near synchronous speed is not possible. This is because at very low rotor frequencies the resistive drop becomes comparable to the applied voltage and any small error in the rotor resistance value gives rise to a substantial error in the rotor flux. Moreover, during problems due to low frequency integration are well known. All these methods using field oriented control involve extensive mathematical computations involving coordinate transformation and parameter estimation. Proposed Solution The proposed methods extends the switching concepts as used in direct torque control of cage rotor induction machines and employs an algorithm for sector changing of the rotor flux vector based on the error in reactive power. This algorithm is inherently position sensorless and does not require any mechanical parameter for flux position estimation. It also provides instantaneous decoupled direct control of the active and reactive power of the machine. The details of the control strategy along with relevant experimental results are presented in the enclosed technical writeup. It is the primary object of invention to invent a novel method for direct control of slip ring induction machine. It is another object of invention to invent a systenn for control of active and a reactive power fronn the rotor side for a grid connected doublyfed induction nnachine without a position encoderIt is another object of invention to invent a systenn for direct control, which is effective, reliable and economical. It is another object of invention to invent a control algorithm, which is embodied in the system so as to execute the method of control of active and reactive power in respect of slip ring induction machine. Further objects of the invention will be clear from the following description The invention will be described in detail with reference to drawings accompanying the specification. The nature of the invention and the manner in which the invention will be performed is clearly described, it is to be noted that an embodiment of invention incorporating novel features oi the invention is described. This description is without limiting the scope oi invention. The statements of the drawings are as follows. Figure 1 shows the schematic diagram of the entire system. Figure 2 shows in block diagram approximate equivalent circuit. Figure 3 shows the phasor diagram of the system. Figure 4a shows the phasor diagram when Irq is varied. Figure 4b shows the phasor diagram when Ird is varied. Figure 5 shows the possible switching states of 3phase voltage inverter Figure 6a shows the orientation of rotor winding in space with respect to which voltage space phasors are drawn. Figure 6b shows the position of voltage space phasors. Figure 7a shows the flux vectors in motoring operation. Figure 7b shows the flux vectors in generating operation. Figure 8 shows the P bana Figure 9 shows the transient response of active power for step change in P* Figure 10 shows stator voltage and current for step charge in active power Figure 11 a shows the stator voltage and current in sub synchronous speed Figure 11b shows the stator voltage and current in super synchronous speed. Figure 12 shows Q and estimated sector during starting on the fly. Figure 13 shows the operation through synchronous speed. Figure 14 shows the switching state, sector and Q sector change. Concept of Direct Power Control The basic concept of direct control of active and reactive power can be appreciated from the phasor diagrams based on the equivalent circuit of the doublyfed machine as shown in fig.2. The stator voltage is ideally constant in magnitude and frequency and hence the stator flux can be assumed to be of constant magnitude laggincj the stator voltage phasor by 90°e. Since the primary objective is independent control of active and reactive powers the stator current is resolved along and perpendicular to the stator voltage phasor to get the active and reactive components respectively. From the phasor diagram in Fig.3 it is noted that the component isq of the stator current has to be controlled to control the active power and isq has to be controlled to control the reactive power. This is achieved in turn by controlling the rotor currents. An injection of irq induces an equivalent but opposite isq as given by the equation. The stator flux magnetising current ims (1) remains constant in this case (since the stator voltage is constant in frequency and magnitude, and the stator resistive drop is negligible), whereby an injection of positive i^^ transfers the reactive power from the stator to the rotor side in accordance with the equation. In conventional fieldoriented control this is accomplished by orienting the rotor currents with respect to the stator flux and controlling them in the synchronous reference frame using PI current loops. For this purpose it is necessary to derive the position of the rotor either by means of a position encoder or by using estimation techniques. The effect of injection of these rotor currents on the airgap and rotor fluxes can be derived by subtracting and adding the respective leakage fluxes. A closer examination of the variation of the rotor flux with variations in the active and reactive power demands is furnished in figs.4 (a) and 4(b). In fig 4(a) the reactive power is fed completely from the stator side i.e. ird = 0. Under this condition if irQ is varied from 0 to full load, the locus of ψt varies along AB, which indicates a predominant change in angle §p between ψs and ψr, whereas the magnitude of ψr does not change appreciably. In other words, a change in the angle 5p would definitely result in a change in the active power in a predictable fashion. For example, in fig.4 (a), which indicates motoring mode of operation, the active power can be increased by decelerating the rotor flux with respect to the stator flux. Conversely it can be reduced by accelerating the rotor flux. In fig. 4(b) the active power demand is maintained constant so that irq is constant and ird is varied from 0 to the rated value of ims . Here the locus of ψr varies along CD resulting in a predominant change in magnitude of mr whereas the variation of §p is small. Therefore, the reactive power drawn from the grid by the stator can be reduced by increasing the nnagnitude of the rotor flux and viceversa. It may be noted that the phasor diagrams as indicated in figs. 4(a) and 4(b) remain the same irrespective of the reference frame; the frequency of the phasor merely change from one reference frame to the other. Therefore, it can be concluded from the above discussion that i. The active power can be controlled by controlling the angular velocity of the rotor flux vector. ii. The reactive power can be controlled by controlling the magnitude of the rotor flux vector. These two basic derivations are used to determine the instantaneous switching state of the rotor side converter to control the active and reactive power as discussed in the following section. Voltage Vectors and their Effects Fig.5 shows the 8 possible switching states of a three phase VSI of which six are active states (S1, S2... S6) and two are zero states (SO, S7), Assuming that the orientation of the three phase rotor winding in space at any instant of time is as given in Fig.6(a), the six active switching states would correspond to the voltage space vectors U1, U2 .... U6 [Fig. 6(b)] at that instant. In order to make an appropriate selection of the voltage vector the space phasor plane is first subdivided into six sectors, I, II, ... VI of 60° angular width. The instantaneous magnitude and angular velocity of the rotor flux space phasor can now be controlled by selecting a particular voltage vector depending on its present location. The effect of the different vectors as reflected on the stator side active and reactive powers, when the rotor flux is positioned in sector 1 is illustrated in the following subsection. A. Effective of Active Vectors on Active Power Considering anti clockwise direction of rotation of the flux vectors in the rotor reference frame to be positive, it may be noted that ψs is ahead of ψr in motoring mode of operation and ψs is behind ψr in generating mode. This is illustrated in fig. 7(a) and fig. 7(b) respectively. In the rotor reference frame the flux vectors rotate in the positive direction at subsynchronous speeds, remain stationary at synchronous speed and start rotating in the negative direction at supersynchronous speeds. In the motoring mode of operation in sector 1, application of voltage vectors U2 and U3 accelerates ψr in the positive direction. This reduces the angular separation between the two fluxes resulting in a reduction of active power drawn by the stator. At subsynchronous speeds, U2 and U3 cause ψr to move in the same anticlockwise direction as ψs, hence the effect on P depends on the difference between the angular velocities of the two fluxes. It may be noted that the factors effecting the angular velocities of the two fluxes ψs and ψr are the slip speed and dc bus voltage respectively. In the rotor reference frames ψs rotates at slip speed and, the rate of change of ψr depends on the dc bus voltage and the applied inverter state. So far a given bus voltage, higher the slip lesser is the relative angular velocity between the two flux vectors, thereby effecting a slower change in P and viceversa. At supersynchronous speed the relative velocity is additive and change in P is faster. In the generating mode of operation, application of vecotrs U2 and U3 result in an increase in angular separation between the two and thereby an increase in the active power generated by the stator, (P being negative for generation, U2 and U3 still result in a reduction of positive active power). The relative speeds of the vectors in subsynchronous and supersynchronous generation are same as in motoring operation; hence the same conclusion can be drawn. Similarly it can be appreciated that the effect of U5 and U6 on the active power would be exactly opposite to that of U2 and U3 in both the motoring and generating modes. Power drawn by the stator being taken an positive and power generated being taken as negative, it may be concluded that, if the rotor flux is in the kth sector, application of vectors U(k + 1) and U(k + 2) would result in an increase in the active power. S. Effect of Active Vectors on Reactive Power From the phasor diagrams Fig 4(a) and Fig. 4(b) it can be seen that the reactive power is dependent on the component of ψr along ψs i.e. ψrd. The angle between ψs and ψr i.e. 5p being small, the magnitude of ψr is approximately equal to ψd, Therefore, when the rotor flux vector is located in sector 1, voltage vectors Uu U2 and Ue increase its magnitude. This holds good irrespective of whether the machine is operating in motoring or generating mode. An increase in magnitude of ψr indicates an increased amount of reactive power being fed from the rotor side and hence, a reduction in the reactive power drawn by the stator resulting in an improved stator power factor. A decrease in magnitude of ψr amounts to lowering of the stator power factor. As a generalization it can, therefore, be said that if the rotor flux resides in the kth, sector, where K = 1, 2, 3, ... 6 switching vectors U(k), U(k+1) and U(k1) reduce the reactive power drawn from the stator side and U(K+2), U(k2, U(k+3) increase the reactive power drawn from the stator side. C Effective of Zero Vector on Active Power The effect of the zero vectors is to stall the rotor flux without effecting its magnitude. This results in an opposite effect on the stator active power in sub"Synchronous and supersynchronous modes of operation. In subsynchronous motoring, application of a zero vector increases 5p as ψs keeps rotating in the positive direction at slip speed. Above the synchronous speed ψs rotates in the counter clockwise direction thereby reducing 8p. Hence positive active power increases for subsynchronous operation and decreases for supersynchronous operation. Active power generated being negative the same conclusion holds true for the generating modes as well. The rate of change of P depends on the slip speed alone as ψr remains stationary in the rotor reference frame. D, Effect of Zero Vector on Reactive Power Since a zero vector does not change the magnitude of the rotor flux its effect on the reactive power is rather small. Nevertheless, there is some small change in Q, its effect being dependent on whether the angle between the stator and rotor fluxes increases or decreases due to the application of a zero vector. An increase in angular separation between the two fluxes reduces ψrd resulting in an increment of Q drawn from the stator side. The converse is true when 5p reduces. It is observed that the change in Q due to the application of (JO or U7 is different in all the 4 modes of operation. This is summarised in Table.1. (The effect on P is also included in this table for the sake of completeness) Control Algorithm With the inferences drawn in the previous section it is possible to switch an appropriate voltage vector in the rotor side at any given instant of time to increase or decrease the active or reactive power in the stator side. Therefore, any given reference for stator active and reactive powers can be trancked within a narrow band by selecting proper switching vectors for the rotor side converter. This is the basis of direct power control strategy. The details of the control algorithm are discussed in the following subsections. A. Measurement of Active and Reactive Power The active and reactive power on the stator side can be directly computed from the stator currents and voltages. Assuming a balanced threephase threewire system, only two currents and two voltages need to be measured. The active and reactive powers can be expressed as Let the references for active and reactive powers be P* and Q* respectively, and the respective allowable bands of excursion of P and 0 on either side of their reference values be Pband and Qband This is illustrated in Fig.8. It is desired that when P crosses P* and hits the upper band the switching vectors which reduce the active power are selected and consequently P is brought down until it hits the lower band. To accomplish this a modified reference P** is defined which is toggled between (P* ^Pband) and (P*  Pband) depending on the sign of (P**  P) as shown in Fig. 8, at instant A, P** is (P* +Pband) and (P*  Pband) is positive. When P crosses P** at instant B, this error becomes negative and instantaneously P** is brought down to (P*  Pband) This continues till instant C when the error again becomes positive and P** is modified to (P* + Pband) This can be formulated as follows. In order to determine the appropriate switching vector at any instant of time, the errors in P and Q and the sector in which the rotor flux vector is presently residing are taken into consideration. Thus the following two switching tables for active vector selection can be generated. Table 2 and 3 correspond to negative Perr and positive Perr respectively If the rotor side converter is switched in accordance to these tables it is possible to control the active and reactive powers in the stator side within the desired error bands. But the use of active vectors alone would result in nonoptimal switching of the converter and also a higher switching frequency. The effect of the zero vectors on P and Q are summarised in Table 1. Since the zero vectors affect both these parameters the usual logic for zero vector selection to enhance / reduce the torque as used in direct torque The choice between SO and S7 is done depending on the minimum inverter switching. For example while switching to a zero vector from S1, SO is selected. On the other hand if the transition to the zero vector is from S2, S7 is selected. Both these transitions then would result in switching of only one arm of the inverter. It may be concluded that these switching strategies would result in close tracking of P* and Q* within the prescribed error bands using nearoptimum switching of the rotor side converter. Sector Identification of Rotor Flux In order to implement the switching algorithm the present sector of the rotor flux has to be identified. The exact position of the rotor flux space phasor is not of importance as far as the selection of the switching vectors is concerned. This is because of the fact that the choice of the rotor voltage vectors is based upon errors in the stator quantities (and not the rotor flux) which are directly measurable. The usual method of computing the rotor flux position and magnitude is rotor voltage integration. This is a fairly standard method and widely used for stator flux estimation in case of direct torque control algorithms for cagerotor induction machine. This method of PWM voltage integration works satisfactorily over a wide frequency range but cannot be used at very low frequencies. The iR drop term becomes comparable to the inverter terminal voltage near zero frequencies and any small error in this term result in an appreciable error in the flux position. This is coupled with integrator saturation problems due to very small offsets at low frequencies. In case of a doublyfed machine controlled from the rotor side it is not possible to implement the rotor voltage integration method near synchronous speed for similar reasons. But in most of the variable speed applications with doublyfed rotorside controlled machines the operating range would be spread on both sides of the synchronous speed to minimize the size of the rotor converters. Therefore, the system should guarantee stable operation at and near synchronous speed and smooth changeover from sub synchronous to super synchronous modes and vice versa. One method of overcoming this problem is to use the rotor position information. If the position of the rotor is known it is possible to transform the stator currents to the rotor coordinates and calculate the rotor flux. This computation would involve machine parameters like Lm and σr. The rotor position information can be directly obtained from an incremental encoder coupled to the machine shaft. Sensorless algorithms can also be employed to estimate the rotor position. Even though these methods of estimation ensure satisfactory operation at and near synchronous speed the computational burden involved is quite high. Moreover, the switching logic does not demand the accurate knowledge of the rotor flux position at all instants of time; the knowledge of the sector in which it is currently positioned is sufficient to choose the appropriate voltage vectors. The proposed method of sector identification in based on the direction of change in Q when a particular switching vector is applied. The concept is illustrated by the following example. Assuming that the present position of the rotor flux is in sector 1, application of switchstates S2 and S6 results in a reduction of Q and application of S3 and S5 results in an increment of 0. When the actual rotor flux vector, if it is moving in the anticlockwise direction (corresponding to sub synchronous operation) crosses over to sector 2, the effect of states S3 and S6 on Q would reverse. Vector U3 would now act to reduce Q instead of increasing it. Similarly the effect of vector U6 on Q would also be opposite. These reversals in the direction of change of Q, when a particular vector is applied can be detected and a decision of sector change may be taken on this basis. Similarly, if the flux vector is rotating in the clockwise direction (super synchronous operation) the effect of states S2 and S5 on Q would change in direction when i/r changes over from Sector 1 to Sector 6. Thus in any particular direction of rotation there are two vectors which can provide the information for sector change. Since the rotor flux vector cannot jump through sectors the change will always be by one sector, either preceding or succeeding. In this method, even though the exact position of the flux is unknown, the sector information can be updated just by observing the changes in Q due to the applied vectors. It may be noted that the effect of the vectors on P would not provide a conclusive inference about the change in sector. The expected direction of change in Q due to the application of any switching vector in the different sectors can be summed up in the following table. It may, however, be noted that in a particular sector not all vectors will be applied. For example, in sector k, vectors U(k) and U(k+3) will never be applied. These vectors would have predominant effect on the reactive power, but their effect on the active power would depend on the actual position of the rotor flux vector in the sector. In most applications there is hardly any requirement of fast transient changes in reactive power so that it is not necessary to apply the strongest vector to effect any change in Q. In the switching logic, therefore, only those vectors are selected which have uniform effects on P and Q in terms of their direction of change irrespective of the position of the rotor flux in a particular sector. For any given vector applied in a particular sector the expected direction of change in Q can be read off Table4. The actual direction of change can be computed from the present value of Q and its previous value. If they are in contradiction then a decision on change of sector is taken. Whether the sector change has to be effected in the clockwise or anticlockwise direction depends on the applied vector and the observed change in Q. This information is stored in another lookup table as furnished below in Tables. To illustrate the algorithm with an example it may be assumed that the rotor flux vector is presently residing in Sector 1 and rotating in the anticlockwise direction (corresponding to sub synchronous speed operation). As long as the flux vector is within the boundary of Sector 1, the direction of change in O will be as expected and the computed direction will match with that stored in Table 4, The quantitative change in Q due to the effect of the vectors will obviously depend on the position of ψr in the sector but the direction of change should be in accordance with this table. In this example, since ψr is rotating in the anticlockwise direction the most widely used active states in Sector 1 will be S2 and S3. When the flux vector has crossed over to Sector 2, S2 will have a more pronounced effect on Q in the same direction as in Sector 1, but the effect of S3 will reverse its direction. The application of S3 will cause a decrement in 0 whereas in Sector 1 it is expected to increase. Hence, the computed direction of change will be opposite to that stored in Table 4. When this is detected the corresponding entry in Table 5 is look at. For Sector 1 and Switch State S3 the entry in Table 5 indicates a positive change in sector. Hence it is updated to Sector 2. Similarly it can be verified that if the flux vectors are rotating in the clockwise direction (corresponding to super synchronous speed) the most commonly used active states will be S6 and S5. When ψr crosses over to Sector 6, S6 will have a predominant effect on Q in the same direction as in Sector 1, but S5 will cause Q to reduce instead of increasing it. This direction of change in 0 is detected from Table 4, and the corresponding entry in Table 5 indicates a change in sector in the negative direction. Hence the sector information is updated from Sector 1 to Sector 6. For reliable detection of the direction of change of Q a minimum switching period of 6 sampling periods (336 microsec) of a particular switching state is maintained. This also puts a maximum switching limit of 4.5 kHz for the rotor side converter. This method of sector identification is independent of any machine parameter but relies on directly measurable fixed frequency quantities. It is also independent of the rotor frequency and can work stable at or near synchronous speed. 4. STARTING Before the rotor side converter is switched on, the entire reactive power is drawn from the stator side. Initially 0* is set to the computed value of 0 after passing it through a lowpass filter of Tip=100 ms. Thereby it is ensured that at the instant of switching the rotor converter, Qen is within Qband and the sector estimation algorithm can be used for correcting to the appropriate sector. Since the minimum switching period on the basis of which a definite decision about sector change can be taken is about 336 microseconds, the algorithm locks onto the correct sector within 1ms even if the actual sector is opposite to the computed sector at switchon. Therefore, the system can be started on the fly without any appreciable transient in rotor or stator currents. Q* is then slowly ramped down to zero (or any other reference) so that the sector updating logic can function properly. It may be noted here that the sector correction logic may give improper inferences for a sudden step change in Q*, however, a transient demand of reactive power is not a practical requirement for the present system, and a gradual change in Q is acceptable. 6, EXPERIMENTAL RESULTS The direct power control algorithm is implemented on a laboratory motorgenerator set. The slipring machine is operated in the generating mode, the prime mover being a dc motor driven by a thyristor drive. The stator of the induction machine is directly connected to 415V ac mains and the rotor circuit is fed via two backtoback IGBT based PWM converters with a common dc bus maintained at 200V. The low dc bus voltage (due to limitation of the present experimental setup) does not allow a wide speedrange; nevertheless the proof of concept can be convincingly established. The control hardware comprises of a TMS320F240 DSP platform operating at 36 MIPs. The 'F240 DSP has two internal ADCs preceded by sampleandhold circuits and two 8channel multiplexers. The present algorithm demands only two stator phase voltages and two stator phase currents to be measured to compute the active and reactive power at the stator terminals. The isolated voltage and current sensor outputs at +/10V level are suitably scaled and shifted to +5V level before being fed to the unipolar ADCs of the processor. The hardware also houses a 4channel external DAC which allows signal monitoring and debugging during development stage. The results presented in the following section are captured from the analog outputs of the DAC channels. The PWM output ports of the processor are programmed in the "forced PWM" mode to generate the switching pulses for the rotor side converter. Transient in active power for a step change in P* from 0 to 0.5 p.u. is shown in Fig.9, Q* is maintained at 0. As P* is changed Perr goes out of the prescribed band. This results in the selection of only the active vectors thereby effecting the fastest possible change in P. The slope of change of P is decided by the rate of change of rotor current, which in turn depends on the dc link voltage. It may be noted from these waveforms that the transient responses in P and Q are perfectly decoupled. The steadystate ripple in P and Q due to switching between the positive and negative error bands can also be observed in Fig.10 illustrates the effect of the active power transient as reflected in the stator current waveform. The steadystate stator voltage and current waveforms for subsynchronous and supersynchronous operation are given in Fig.11 (a) and Fig.11 (b) respectively. With Q* being set to 0 the reactive power for the machine is supplied fully from the rotor side resulting in upf generation. This is particularly important for windpower generating systems. Under low wind condition the active power generation is poor; whereas all the generators connected to the grid put together demand a huge amount of reactive power resulting in a very poor power factor of the wind farm. One of the important requirements of windpower generators is that the machines have to be "cutin" when the turbine speed crosses a given limit. The method of "onthefly" starting has been discussed in Section VI. The relevant waveforms of Q and the computed sector are given in Fig.12. (The sector information is scaled and output through DAC such that the analog output voltage is IV multiplied by the sector number). Before the rotor converter is switched on, the sector information as can be seen from the plot is erroneous. However, the computed sector locks onto the actual sector instantaneously as the rotor circuit is excited. 0 is gradually ramped down to zero. Fig. 13 gives the rotor current waveform along with the sector information for transition through synchronous speed. During sub synchronous speed operation the flux vectors rotate in the anticlockwise direction in the rotor reference frame; hence the sector number increases from 1 to 6 and resets back to 1. This is represented by the ascending staircase waveform. As the rotor moves over to super synchronous speed the flux vectors start rotating in the clockwise direction. Therefore the sector number changes in the reverse order as seen by the descending staircase. The changeover from sub synchronous to super synchronous speed is observed to be smooth without any transients in P and Q. The instant of sector change from Sector 1 to Sector 2 is shown in a magnified scale in Fig. 14. It is noted that with the application of switching state S3 in Sector 1, Q decreases instead of increasing. S3 is applied for 6 sampling periods (336 us) at the end of which the decision for sector change is taken. As the sector is updated to 2, the switching state now selected is S4 which effects fast increment in 0 as desired. Salient Feature of Invention A method of direct power control for grid connected doublyfedslipring induction machine is presented. The stator active and reactive powers are controlled within hysteresis bands by adopting a switching algorithm on the rotor side, it is proposed that instead of estimating the exact position of the rotor flux, the information of the sector in which it resides is sufficient for switching the correct inverter state. A novel method for sector identification based on the direction of change of reactive power is presented. The control algorithm uses only stator quantities for active and reactive power measurement and is inherently position sensorless. It is computationally simple and does not incorporate any machine parameter. Relevant experimental results to validate the concept are presented. The direct power control method can be an attractive proposition for slipring induction generators in windenergy application. We claim 1. A method for direct control of active and reactive power from rotor side for a grid connected doubly fed slip ring induction machine without position encoder comprising extension of switching concepts as used in direct control of cage rotor induction machine, employing an algorithm for sector changing of the rotor flux vector based on the error in reactive power; the algorithm is inherently position sensorless and does not require any parameter for flux position estimation, the said algorithm further provides instantaneous de coupled direct control of active and reactive power of the machine. 2. A system for direct control of active and reactive power from rotor side for a grid connected doubly fed slip ring induction machine without position encoder, the said system comprises an induction motor the stator of the said induction motor is energised by a three phase power supply, the rotor of the induction being energised by a circuit comprising a three phase transformer connected in series to a three phase front end converter, which is connected in series to a three phase inverter, the three phase inverter is connected to the rotor of the said induction motor wherein a capacitor is connected between three phase converter and three phase inverter. 3. An algorithm developed for the system for direct control of active and reactive power from rotor side for a grid connected doubly fed slip ring induction machine without position encoder. 

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Patent Number  212361  

Indian Patent Application Number  797/MAS/1999  
PG Journal Number  07/2008  
Publication Date  15Feb2008  
Grant Date  03Dec2007  
Date of Filing  06Aug1999  
Name of Patentee  INDIAN INSTITUTE OF SCIENCE  
Applicant Address  BANGALORE  560 012,  
Inventors:


PCT International Classification Number  H02J 3/18  
PCT International Application Number  N/A  
PCT International Filing date  
PCT Conventions:
