Title of Invention	"A METHOD OF ESTEMATING A CHANNEL IN A WIRELESS COMMUNICATION SYSTEM AND AN APPARATUS THEREOF"
Abstract	A method of estimating a channel in a wireless communication system in a processor (1130, 1170), wherein the estimation of a channel in a wireless communication system is characterized by: obtaining at least two groups of received pilot symbols for at least two sets of pilot subbands, one group of received pilot symbols for each set of pilot subbands, wherein the second group is staggered with respect to the first group; dividing received pilot symbols into even symbols and odd symbols, wherein the even symbols include actual and excess components and the odd symbols include actual and excess components; determining an even effective estimate and an odd effective estimate based on the even symbols and the odd symbols, respectively; selecting a first set of time-filter coefficients for estimating an actual channel; selecting a second set of time-filter coefficients for estimating an excess channel; time-filtering said actual channel based on at least the even effective estimate, the odd effective estimate, the first set of time-filter coefficients and the second set of time-filter coefficients; and time-filtering said excess channel based on at least the even effective estimate, the odd effective estimate, the first set of time-filter coefficients and the second set of time-filter coefficients, wherein said actual component include data at or within a prefix length, and excess components include data outside of the prefix length. FIG. 2A

Title of Invention

"A METHOD OF ESTEMATING A CHANNEL IN A WIRELESS COMMUNICATION SYSTEM AND AN APPARATUS THEREOF"

Abstract

A method of estimating a channel in a wireless communication system in a processor (1130, 1170), wherein the estimation of a channel in a wireless communication system is characterized by: obtaining at least two groups of received pilot symbols for at least two sets of pilot subbands, one group of received pilot symbols for each set of pilot subbands, wherein the second group is staggered with respect to the first group; dividing received pilot symbols into even symbols and odd symbols, wherein the even symbols include actual and excess components and the odd symbols include actual and excess components; determining an even effective estimate and an odd effective estimate based on the even symbols and the odd symbols, respectively; selecting a first set of time-filter coefficients for estimating an actual channel; selecting a second set of time-filter coefficients for estimating an excess channel; time-filtering said actual channel based on at least the even effective estimate, the odd effective estimate, the first set of time-filter coefficients and the second set of time-filter coefficients; and time-filtering said excess channel based on at least the even effective estimate, the odd effective estimate, the first set of time-filter coefficients and the second set of time-filter coefficients, wherein said actual component include data at or within a prefix length, and excess components include data outside of the prefix length. FIG. 2A

Full Text	The present invention relates to a method of estimating a channel in a wireless communication system and an apparatus thereof. [0002] The present invention relates generally to data communication, and more specifically to time filtering for excess delay mitigation in orthogonal frequency division multiplexing (OFDM) systems. II. Background [0003] OFDM is a multi-carrier modulation technique that effectively partitions the overall system bandwidth into multiple (NF) orthogonal subbands. These subbands are also referred to as tones, subcarriers, bins, and frequency channels. With OFDM, each subband is associated with a respective subcarrier that may be modulated with data. Up to NF modulation symbols may be transmitted on the NF subbands in each OFDM symbol period. Prior to transmission, these modulation symbols are transformed to the time-domain using an NF-point inverse fast Fourier transform (IFFT) to obtain a "transformed" symbol that contains NF chips. [0004] OFDM can be used to combat frequency selective fading, which is characterized by different channel gains at different frequencies of the overall system bandwidth. It is well known that frequency selective fading causes intersymbol interference (ISI), which is a phenomenon whereby each symbol in a received signal acts as distortion to one or more subsequent symbols in the received signal. The ISI distortion degrades performance by impacting the ability to correctly detect the received symbols. Frequency selective fading can be conveniently combated with OFDM by repeating a portion of each transformed symbol to form a corresponding OFDM symbol. The repeated portion is commonly referred to as a cyclic prefix. [0005] The length of the cyclic prefix (i.e., the amount to repeat for each OFDM symbol) is dependent on delay spread. The delay spread of a wireless channel is the time span or duration of an impulse response for the wireless channel. This delay spread is also the difference between the earliest and latest arriving signal instances (or multipaths) at a receiver for a signal transmitted via the wireless channel by a transmitter. The delay spread of an OFDM system is the maximum expected delay spread of the wireless channels for all transmitters and receivers in the system. To allow all receivers in the system to combat ISI, the cyclic prefix length should be equal to or longer than the maximum expected delay spread. However, since the cyclic prefix represents an overhead for each OFDM symbol, it is desirable to have the cyclic prefix length be as short as possible to minimize overhead. As a compromise, the cyclic prefix length is typically selected such that the cyclic prefix contains a significant portion of all multipath energies for most receivers in the system. [0006] An OFDM system can withstand a delay spread that is smaller than or equal to the cyclic prefix length. When this is the case, the NF subbands are orthogonal to one another. However, a given receiver in the system may observe excess delay spread, which is a delay spread that is greater than the cyclic prefix length. Excess delay spread can cause various deleterious effects, such as ISI and channel estimation errors, both of which can degrade system performance as described below. There is therefore a need in the art for techniques to mitigate the deleterious effects of excess delay spread in an OFDM system. SUMMARY [0007] Techniques for transmitting pilot and estimating the response of a wireless channel with excess delay spread are described herein. [0008] In an aspect, a method of estimating a channel in a wireless communication system comprises obtaining at least two groups of received pilot symbols for at least two sets of pilot subbands, one group of received pilot symbols for each set of pilot subbands, wherein a second group is staggered with respect to a first group, dividing received pilot symbols into even symbols and odd symbols, wherein the even symbols include actual and excess components and the odd symbols include actual and excess components, determining an even effective estimate and an odd effective estimate based on the even symbols and the odd symbols, respectively, selecting a first set of time-filter coefficients for estimating an actual channel, selecting a second set of time-filter coefficients for estimating an excess channel, time-filtering for an actual channel based on at least the even effective estimate, the odd effective estimate, the first set of time-filter coefficients and the second set of time-filter coefficients, and time-filtering for an excess channel based on at least the even effective estimate, the odd effective estimate, the first set of time-filter coefficients and the second set of time-filter coefficients. [0009] In another aspect an apparatus in a wireless communication system comprising means for obtaining at least two groups of received pilot symbols for at least two sets of pilot subbands, one group of received pilot symbols for each set of pilot subbands, wherein a second group is staggered with respect to a first group, means for dividing received pilot symbols into even symbols and odd symbols, wherein the even symbols include actual and excess components and the odd symbols include actual and excess components, means for determining an even effective estimate and an odd effective estimate based on the even symbols and the odd symbols, respectively, means for selecting a first set of time-filter coefficients for estimating an actual channel, means for selecting a second set of time-filter coefficients for estimating an excess channel, means for time-filtering for an actual channel based on at least the even effective estimate, the odd effective estimate, the first set of time-filter coefficients and the second set of time-filter coefficients, and means for time-filtering for an excess channel based on at least the even effective estimate, the odd effective estimate, the first set of time-filter coefficients and the second set of time-filter coefficients. [0010] In yet another aspect, a computer readable media embodying a method for estimating a channel in a wireless communication system comprising obtaining at least two groups of received pilot symbols for at least two sets of pilot subbands, one group of received pilot symbols for each set of pilot subbands, wherein a second group is staggered with respect to a first group, dividing received pilot symbols into even symbols and odd symbols, wherein the even symbols include actual and excess components and the odd symbols include actual and excess components, determining an even effective estimate and an odd effective estimate based on the even symbols and the odd symbols, respectively, selecting a first set of time-filter coefficients for estimating an actual channel, selecting a second set of time-filter coefficients for estimating an excess channel, time-filtering for an actual channel based on at least the even effective estimate, the odd effective estimate, the first set of time-filter coefficients and the second set of time-filter coefficients, and time-filtering for an excess channel based on at least the even effective estimate, the odd effective estimate, the first set of time-filter coefficients and the second set of time-filter coefficients. [0011] Various aspects and embodiments of the invention are described in further detail below. BRIEF DESCRIPTION OF THE DRAWINGS [0012] The features and nature of the present invention will become more apparent from the detailed description set forth below when taken in conjunction with the drawings in which like reference characters identify correspondingly throughout and wherein: [0013] ' FIG. 1 shows an OFDM modulator for an OFDM system; [0014] FIGS. 2A and 2D show a wireless channel with excess delay spread and its effective channel, respectively; [0015] FIGS. 2B and 2C show a sequence of received chips for the wireless channel; [0016] FIG. 3 shows a subband structure that may be used for the OFDM system; [0017] FIG. 4 shows an access point and a terminal in the OFDM system; and [0018] FIG. 5 shows a channel estimator. FIG. 6 shows a terminal in a wireless communication system. FIG. 7 shows a flowchart for a method of estimating a channel in a wireless communication system. [0019] The word "exemplary" is used herein to mean "serving as an example, instance, or illustration." Any embodiment or design described herein as "exemplary" is not necessarily to be construed as preferred or advantageous over other embodiments or designs. [0020] The techniques described herein for time filtering for excess delay mitigation may be used for various communication systems such as an orthogonal frequency division multiplexing (OFDM)-based system, an Orthogonal Frequency Division Multiple Access (OFDMA) system, a Code Division Multiple Access (CDMA) system, a Time Division Multiple Access (TDMA) system, a Frequency Division Multiple Access (FDMA) system, a single-input single-output (SISO) system, a multiple-input multiple-output (MIMO) system, and so on. [0021] In an OFDM system, a cyclic prefix is inserted at the beginning of each OFDM symbol to remove interference across successive symbols. "When the delay spread of the channel is less than the cyclic prefix and the receiver is synchronized to choose the appropriate EFT window, there is no inter-symbol interference (ISI) between successive OFDM symbols. Further, linear convolution with the channel impulse response is converted to a circular convolution,- and the orthogonality of the carriers is preserved. In other words, there is no inter-carrier interference (ICI) between different carriers within the same OFDM symbol. [0022] When the delay spread of the channel exceeds the cyclic prefix, there is ICI as well as ISI, and this could degrade the performance of the OFDM system. Increasing the length of the cyclic prefix to avoid this degradation could lead to an unacceptable overhead in the- system. .. In addition to the introduction of ICI/TSI, the presence of excess delay spread could lead to further degradation in a coherent receiver that needs to estimate the channel. Specifically, if the number of channel taps has increased and the pilot resources allocated for channel estimation: could be insufficient. Clearly, the degradation in such a scenario would depend on the allocated resources as well as the amount of excess delay spread. [0023] As with the cyclic prefix, increasing the resources for channel estimation may lead to an unacceptable increase in overhead. Degradation in channel estimation could be quite significant in some scenarios of practical interest, overshadowing the intrinsic degradation due to ICI and ISI. Using channel estimation techniques that account for the presence of excess delay spread mitigate such effects. [0024] FIG. 1 shows a block diagram of an OFDM modulator 100 for an OFDM system. The data to be transmitted is typically encoded and interleaved to generate code bits, which are then mapped to modulation symbols. The symbol mapping is performed by (1) grouping the code bits into B-bit binary values, where B ≥ 1, and (2) mapping each B-bit value to a specific, modulation symbol based on a modulation scheme (e.g., M-PSK or M-QAM, where M = 2B). Each modulation symbol is a complex value in a signal constellation corresponding to the modulation scheme. For each OFDM symbol period, one "transmit" symbol is sent on each of the NF subbands. Each transmit symbol can be either a modulation symbol for pilot/data or a signal value of zero (i.e., a "zero symbol"). An IFFT unit 110 performs an NFpornt IFFT on the NF transmit symbols for the NF total subbands in each OFDM symbol period and provides a transformed symbol that contains NF chips. The IFFT may be expressed as: (Equation Removed) where S is an NF x1 vector of transmit symbols for the NF subbands; WNFNF IS an NF xNF discrete Fourier transform (DFT) matrix; s is an NF xl vector of time-domain chips; and "H " denotes the conjugate transpose. The DFT matrix WNFXNF: is defined such that the (n,m) -th entry, wn,m, is given as: (Equation Removed) where n is a row index and m is a column index. WN xNp is an inverse DFT matrix. [0025] A cyclic prefix generator 120 repeats a portion of each transformed symbol to obtain a corresponding OFDM symbol that contains Nc chips, where Nc = NF + Ncp and Ncp is the cyclic prefix length. An OFDM symbol period is the duration of one OFDM symbol, which is Nc chip periods. The chips are conditioned and transmitted via a wireless channel. [0026] FIG. 2A shows an exemplary impulse response 210 of a wireless channel with excess delay spread. Channel impulse response 210 includes two taps 212 and 214 for two multipaths in the wireless channel. Actual tap 212 has a complex gain of h1 and is located at tap index 1. Excess tap 214 has a complex gain of h, and is located at tap index Ne, which is outside of the cyclic prefix length Ncp. As used herein, "main channel" refers to the portion of the channel impulse response that is at or within the cyclic prefix length, "excess channel" refers to the portion of the channel impulse response that is outside of the cyclic prefix length, and "excess" refers to the difference between the tap index of an excess channel tap and the cyclic prefix length. For channel impulse response 210, the main channel includes one actual tap 212, the excess channel includes one excess tap 214, and the excess for tap 214 is Nex = Ne- Ncp. [0027] FIG. 2B shows a sequence 220 of received chips for the wireless channel shown in FIG. 2A. Received chip sequence 220 is a convolution of a transmitted chip sequence with taps 212 and 214 for the wireless channel. Received chip sequence 220 is composed of (1) a chip sequence 222 generated by convolving main channel tap 212 with the transmitted chip sequence and (2) a chip sequence 224 generated by convolving excess channel tap 214 with the transmitted chip sequence, where Si,- denotes the i-th chip for the current OFDM symbol, xi denotes the i-th chip for the previous OFDM symbol, and i = 1 .. Nc. [0028] FIG. 2C shows the decomposition of received chip sequence 220 into different components. Chip sequence 224 in FIG. 2B is replaced with (1) a chip sequence 226 generated by a circular convolution of excess channel tap 214 with the Nc chips for the current OFDM symbol, (2) a chip sequence 228 for the tail end of the previous OFDM symbol, and (3) a chip sequence 230 for the tail end of the current OFDM symbol. Chip sequences 222 and 226 represent the sequences that would have been received for taps 212 and 214 if the cyclic prefix length were sufficiently long and tap 214 is part of the main channel. However, since this is not the case, chip sequences 228 and 230 are both due to the excess delay spread. Chip sequence 228 represents the leakage of the previous OFDM symbol into the current OFDM symbol and is the source of intersymbol interference. Chip sequence 230 represents the disturbance to the circular convolution and is the source of intercarrier interference (ICI) and channel attenuation. [0029] The intersymbol interference observed in each subband may be expressed as: r where X is an NF x1 vector of transmit symbols for the previous OFDM symbol; WH NexNF is an Nex xNF matrix with the last Nex rows of WHNFXNF ; and W1xNex (k) is a 1 x Nex vector with the first Nex elements of the k-th row of WNF xN. The operation WHNEXNF X generates an Nex X1 vector XNex that contains the last Nex chips of the previous OFDM symbol. The multiplication of XN with W1xNex (k) generates the interference due to these last Nex chips on subband k. [0030] The noise power on each subband due to intersymbol interference can be expressed as: (Equation Removed) where Es is the transmit symbol energy, he2 is the power of the excess channel, and σ 2ISI, is the noise power due to ISI on each subband. As shown in equation (4), the ISI noise power per subband is (1) proportional to the excess channel energy he2, (2) proportional to the excess Nex, which is indicative of the amount of leakage of the previous OFDM symbol onto the current OFDM symbol, and (3) inversely related to the number of total subbands since the total ISI noise power is distributed over the NF subbands. [0031] The noise power on each subband due to intercarrier interference can be computed in similar manner as for intersymbol interference and expressed as: (Equation Removed) where σ2ICI is the noise power due to ICI on each subband. [0032] FIG. 2D shows an "effective" channel 240 for the wireless channel shown in FIG. 2A. Referring back to FIG. 2C, chip sequence 226 represents the contribution due to excess channel tap 214 (assuming that the cyclic prefix is long enough), and chip sequence 230 represents the source of ICI due to the excess channel. The subtraction operation for chip sequence 230 results partly in a reduction of the signal power for each subband. This subtraction can be accounted for by scaling down excess channel tap 214 by a factor of (1-Nex/NF). As shown in FIG. 2D, effective channel 240 includes tap 212 having the complex gain of h1 and a tap 216 having a complex gain of hc (1 - Nex /NF). The reduction in the gain of tap 216 relative to the gain of tap 214 is referred to as "channel attenuation" and results from excess delay spread for tap 214. The amount of attenuation is related to the excess Nex. [0033] A receiver performs channel estimation in order to derive a channel estimate for the wireless channel. Channel estimation is typically performed based on pilot symbols, which are modulation symbols that are known a priori by the receiver. The pilot symbols may be transmitted in various manners as described below. [0034] FIG. 3 shows an exemplary subband structure that may be used for the OFDM system. The OFDM system has an overall system bandwidth of BW MHz, which is partitioned into NF orthogonal subbands using OFDM. Each subband has a bandwidth of BW/NF MHz. For a spectrally shaped OFDM system, only NU of the NF total subbands are used for data/pilot transmission, where NU to allow the system to meet spectral mask requirements. For simplicity, the following description assumes that all NF subbands may be used in the OFDM system. [0035] For the sake of illustration, an OFDM system is considered where channel estimation is based on -uniformly spaced pilots in the frequency domain. The kth received OFDM symbol in the frequency domain can be written as (Equation Removed) where • P is the number of pilots carriers, and D is the number of channel taps assumed by the receiver. • the vectors Y,H, w are of length P and the noise w is white complex Gaussian with variance N0. ■ the matrix WP,D is the PxD submatrix of the unnorinalized DFT matrix (Equation Removed) where N is the total number of snbcarriers. [0036] The number of channel taps D ≤ P. However, in an embodiment a longer channel estimate for dealing with scenarios where the channel has a delay spread larger than the cyclic prefix. To get a longer channel estimate, pilots are staggered across successive OFDM symbols, i.e., the pilot carrier indices are changed in successive OFDM symbols as described below. Staggered pilots • [0037] For simplicity, assuming a two symbol staggering pattern: if the uniformly spaced pilot carriers are of the formin the even symbols, they would be in the odd symbols. "With such staggering, we can get an estimate of up to a length 2P by using the pilot observations from two neighboring OFDM symbols. Specifically, assume a channel with 2P time domain taps. Then [0038] For farther simplicity, set , so that the staggering is between phases 0 and. though the above expression can be carried through for any . We then have (Equation Removed) [0039] Similarly, for the odd symbols, (Equation Removed) [0040] Thus, the pilot observations in the even and odd symbols can be written as (Equation Removed) where and the superscripts "a" and "e" denote the "actual"'and "excess" taps that correspond to ' and ' respectively. Actual tap and excess tap are as discussed above and as shown in, for example, FIG. 2A, where actual tap 212 and excess tap 214 are shown. "Excess" refers to the difference between the tap index of an excess channel tap and the cyclic prefix length. For a channel impulse response, the main channel includes one actual tap and the excess channel includes one excess tap, and the excess for the excess tap is Nex = Ne - Ncp. [0041] To get an estimate of the channel from the observations in Equation 7, one possibility is to use a least-squares approach to estimate the effective time-domain channel. Equation 8 shows an even effective estimate and an odd effective estimate: (Equation Removed) [0042] The effective estimates above include both actual and excess components. A simple way to get the full 2P tap channel estimate is (Equation Removed) [0043] Equation 9 is just a special case of a more general operation where the time- domain estimates in Equation 8 (obtained every OFDM symbol) are averaged across multiple OFDM symbols. Such averaging is referred to as time-filtering, and it is done separately for each individual time-domain tap. The resulting estimate of tap l at any OFDM symbol m (odd or even) can be written as (Equation Removed) where Nf and Nb are the number of non-causal and causal taps, respectively. In this framework, Equation 9 corresponds to choosing Nf = 0, Nb = 2 and (Equation Removed) [0044] Thus, one set of time-filter coefficients is chosen for estimating the actual channel (l [0045] Consider more general strategies for choosing the time-filter coefficients for the two halves. For clarity, the filter co-efficients for l filter co-efficients for l ≥ P is denoted by ßn. Time-filtering for the actual channel [0046] Apart from separating the actual and excess channels, the choice of time-filter coefficients are governed by other constraints as well. Time-filtering enables the capture of additional pilot energy and improvement in the reliability of channel estimates. However, using a long time-filter can introduce degradations due to time-variations of the channel. [0047] For the sake of illustration, focus on the observed lth time-domain channel tap in an even OFDM symbol, and assume that channel varies linearly over the Nf +Nb symbols that are used by the time filter. Using Equation 8, we have (Equation Removed) where δa and δe are the slopes of the actual and excess channels at tap I. Ideally, these time-variations would be canceled along with the excess channel. Hence, the constraints on the time-filter co-efficients can be summarized as: (Equation Removed) [0048] Since these constraints are invariant to a scale factor in the co-efficients, a normalization constraint may be imposed, e.g. that the channel estimate be unbiased, which means (Equation Removed) [0049] For example, given a three tap filter with one non-causal tap, i.s..Nf = 1,Nb =2, and the constraints in Equation 10 and Equation 11, the solution is {0.25,0.5,0.25]. In the absence of excess channel taps, the optimal solution would be {0.33,0.33,0.33}. [0050] When the number of coefficients is greater than the number of linearly independent constraints, the coefficients can be chosen to minimize the noise variance in the time-filtered estimate, i.e., (Equation Removed) under the constraints of Equation 10 and Equation 11. It would be apparent to those skilled in the art that since the constraints are linear and the objective function is quadratic, this optimization can be solved using Lagrange multiplier techniques. Time-filtering for the excess channel [0051] Thus far, selection of filter taps has been restricted to the first P taps. For l > P, the taps correspond to the excess channel, and are denoted by{ßn}. [0052] In choosing{ßn}, the goal is reversed from that for l excess taps are kept and contributions from the first P taps are eliminated. Hence, the constraints in Equation 10 are modified as: (Equation Removed) [0053] Only the first constraint has changed, and a scale factor constraint as in Equation 11 can be imposed. For the three tap non-causal filter, the solution for {B-1,,ß0,ß1}is {-0.25,0.5,-0.25}. It would be apparent to those skilled in the art that similar solutions can be obtained for other filter lengths (and other staggering patterns) as well. Efficient generation of frequency domain estimate [0054] In a modem implementation, the channel estimate in the frequency domain is finally obtained on a per-interlace basis. That is, to reduce the number of computations involved in the FFT operation to get the channel estimates in the frequency domain, a P pt FFT is performed on the time domain channel estimate (after introducing a suitable phase ramp), thereby resulting in the channel estimates for the interlace of interest. With the estimation of the channel taps corresponding to the excess delay in the channel, there are 2P taps for the channel estimate in the time domain. A channel estimate for the required interlace can be obtained with a single P pt FFT operation. In particular, let the IP channel taps in the time domain be represented by h = [ha he] where ha and he are each P length vectors. Given the frequency estimate for the P subcarriers {d = 0,1,2, P-1) corresponding to the interlace m (m = 0,l,...7), then the frequency domain channel estimate for the d"' carrier in mth interlace is given by (Equation Removed) [0055] The extra P taps of the channel result in some trivial complex multiplications (in four out of eight values of m) and additions. The phase ramp operation followed by the P pt EFT would have been performed irrespective of the number of channel taps being P. However, not truncating the channel to P taps, thereby allowing the extra P taps, requires additional memory for the buffering purposes. [0056] Several assumptions and imposed limitations in the above discussion were made for the sake of illustration. Specifically, [0057] Staggering pattern: A simplistic staggering pattern with just two phases (0 and 4) was assumed. It would be apparent to those skilled in the art that the disclosed embodiments generalize to any other staggering pattern across different OFDM symbols. In each symbol, the pilots are uniformly spaced so that the excess channel aliases in the time-domain. The choice of the staggering pattern could be based on other factors and is of interest in itself. [0058] Least-squares criterion: In going from the pilots in the frequency domain to the aliased time domain channel estimate, a least-squares approach is used, which translates to an IFFT. It would be apparent to those skilled in the art that other criteria for deriving the time-domain estimate are possible, e.g. an MMSE approach. [0059] A key point here is the relationship between the time-domain channel and pilot observations that is induced by staggering. See Equation 7. [0060] Time-filter length: A three tap filter for illustration was assumed. Clearly, the approach is applicable for any number of taps that is greater than two, and the filter can total number of pilot observations is greater than the total channel length assumed, otherwise, perfect estimation of the complete channel is not possible. [0061] Filter co-efficient selection: In choosing the filter co-efficients in accordance with an embodiment, it is assumed that the same set of co-efficients are used for all the taps in the actual channel, and a different set is used for all the taps in the excess channel. In another embodiment, a different set of co-efficients is used for each tap in the actual channel as well as each tap in the excess channel (resulting in 2P sets of filter co-efficients in the example). Additional constraints have been imposed that the lime-variation of the channel must be cancelled or suppressed when choosing the co- • efficients. These constraints can be released depending on the number of time-filter co efficients or other system design requirements. [0062] Linear variation model: Finally, in formulating the constraints in Equation 10 etc, a model has been used where the channel varies linearly over the duration of interest. Other approaches can be used to derive the constraints, e.g. a statistical model can be assumed for the channel correlation over time and the problem can be posed in terms of minimizing the variance of the time-variation errors. [0063] For clarity, the pilot transmission and channel" estimation techniques have been described for an OFDM system. These techniques may be used for other multi-carrier modulation techniques such as discrete multi tone (DMT). [0064] FIG. 4 shows a block diagram of an access point 1100 and a terminal 1150 in the OFDM system. On the downlink, at access point 1100, a transmit (TX) data processor 1110 receives, formats, codes, interleaves, and modulates (i.e., symbol maps) traffic data and provides modulation symbols (or simply, "data symbols"). An OFDM modulator 1120 receives the data symbols and pilot symbols, performs OFDM modulation as described for FIG. 1, and provides a stream of OFDM symbols. Pilot symbols are transmitted in a staggered manner. A transmitter unit (TMTR) 1122 receives and converts the stream of OFDM symbols into one or more analog signals, conditions (e.g., amplifies, filters, and frequency upconverts) the analog signals to generate a downlink signal, and transmits the signal via an antenna 1124 to the terminals. [0065] At terminal 1150, an antenna 1152 receives the downlink'signal and provides a received signal to a receiver unit (RCVR) 1154. Receiver unit 1154 conditions (e.g., filters, amplifies, and frequency downconverts) the received signal, digitizes the conditioned signal, and provides received chips to an OFDM demodulator 1156. [0066] FIG. 5 shows an embodiment of OFDM demodulator 1156. A cyclic prefix removal unit 1212 removes the cyclic prefix appended to each OFDM symbol. An FFT unit 1214 then transforms each received transformed symbol to the frequency domain using an NF-point FFT and obtains NF received symbols for the NF subbands. FFT unit 1214 provides received pilot symbols to a processor 1170 and received data symbols to a detector 1216. Detector 1216 further receives a frequency response estimate Hm ,d(k) for the downlink from processor 1170, performs detection on the received data symbols WE CLAIM: 1. A method of estimating a channel in a wireless communication system in a processor (1 130, 1170), wherein the estimation of a channel in a wireless communication system is characterized by: obtaining at least two groups of received pilot symbols for at least two sets of pilot subbands, one group of received pilot symbols for each set of pilot subbands, wherein the second group is staggered with respect to the first group; dividing received pilot symbols into even symbols and odd symbols, wherein the even symbols include actual and excess components and the odd symbols include actual and excess components; determining an even effective estimate and an odd effective estimate based on the even symbols and the odd symbols, respectively; selecting a first set of time-filter coefficients for estimating an actual channel; selecting a second set of time-filter coefficients for estimating an excess channel; time-filtering said actual channel based on at least the even effective estimate, the odd effective estimate, the first set of time-filter coefficients and the second set of time-filter coefficients; and time-filtering said excess channel based on at least the even effective estimate, the odd effective estimate, the first set of time-filter coefficients and the second set of time-filter coefficients, wherein said actual component include data at or within a prefix length, and excess components include data outside of the prefix length. 2. The method as claimed in claim 1, wherein said dividing further comprises uniformly spacing the received pilot symbols of the second group from the received pilot symbols of the first group. 3. The method as claimed in claim 2, wherein said dividing further comprises staggering the at least two groups of received pilot symbols according to a two symbol staggering pattern. 4. The method as claimed in claim 2, wherein said dividing M e r comprises staggering the at least two groups of received pilot symbols according to an n-symbol (n>) staggering pattern. 5. An apparatus in a wireless communication system, the apparatus characterized by: means for obtaining at least two groups of received pilot symbols for at least two sets of pilot subbands, one group of received pilot symbols for each set of pilot subbands, wherein the second group is staggered with respect to the first group; means for dividing received pilot symbols into even symbols and odd symbols, wherein the even symbols include actual and excess components and the odd symbols include actual and excess components; means for determining an even effective estimate and an odd effective estimate based on the even symbols and the odd symbols, respectively; means for selecting a first set of time-filter coefficients for estimating an actual channel; means for selecting a second set of time-filter coefficients for estimating an excess channel; means for time-filtering said actual channel based on at least the even effective estimate, the odd effective estimate, the first set of time-filter coefficients and the second set of time-filter coefficients; and means for time-filtering said excess channel based on at least the even effective estimate, the odd effective estimate, the first set of time-filter coefficients and the second set of time-filter coefficients, wherein said actual components include data at or within a prefix length, and excess components include data outside of the prefix length. 6. An apparatus for estimating a channel in a wireless communication system, the apparatus comprising: a memory unit (1 132, 1 172); a processor (1 130, 1 170) coupled to the memory unit (1 132, 1 172), wherein the processor is configured to perform steps being characterized by: obtaining at least two groups of received pilot symbols for at least two sets of pilot subbands, one group of received pilot symbols for each set of pilot subbands, wherein the second group is staggered with respect to the first group; dividing said received pilot symbols into even symbols and odd symbols, wherein the even symbols include actual and excess components and the odd symbols include actual and excess components; determining an even effective estimate and an odd effective estimate based on the even symbols and the odd symbols, respectively; selecting a first set of time-filter coefficients for estimating an actual channel; selecting a second set of time-filter coefficients for estimating an excess channel; time-filtering said actual channel based on at least the even effective estimate, the odd effective estimate, the first set of time-filter coefficients and the second set of time-filter coefficients; and time-filtering said excess channel based on at least the even effective estimate, the odd effective estimate, the first set of time-filter coefficients and the second set of time-filter coefficients, wherein said actual components include data at or within a prefix length, and excess components include data outside of the prefix length. 7. A communication signal demodulating apparatus comprising an OFDM demodulator (1 156) configured to receive data comprising a plurality of symbols, the plurality of symbols comprising pilot symbols and data symbols, the OFDM demodulator (1 156) being characterized by: a cyclic prefix removal unit (1 2 12) for removing a cyclic prefix appended to said plurality of symbols; a Fast Fourier Transform unit (1214) for transforming said plurality of symbols into the frequency domain; and a detector (1 2 16); and a processor (1 170) configured to receive said pilot symbols from the Fast Fourier Transform unit (1 2 14) and perform channel estimation on said received pilot symbols by dividing said received pilot symbols into even symbols and odd symbols, wherein the even symbols include actual and excess components and the odd symbols include actual and excess components, and determining an even effective estimate and an odd effective estimate based on the even symbols and the odd symbols, respectively, the processor (1170) further configured to determine at least one frequency response value based on the even effective estimate and the odd effective estimate, wherein the detector (1216) is configured to receive said data symbols from the Fast Fourier Transform unit (1214) and the at least one frequency response value from the processor (1 170) to obtain estimates of transmitted data symbols and provide detected symbols, and wherein said actual components include data at or within a prefix length, and said excess components include data outside of the prefix length. 8. The communication signal demodulation apparatus as claimed in claim 7, wherein the processor (1 170) comprises a channel estimator (1220) configured to receive pilot symbols and perform channel estimation. 9. The communication signal demodulating apparatus as claimed in claim 8, wherein the channel estimator (1220) comprises a pilot detector (1222) having an output operably connected to an input of an inverse Fast Fourier Transform unit (1224) having an output operably connected to an input of a repetition processor (1226) having an output operably connected to an input of a combinerlfilter (1228) having an output operably connected to an input of a threshold and padding processor (1230) having an output operably connected to an input of an estimator Fast Fourier Transform unit (1 232). 10. The communication signal demodulating apparatus as claimed in claim 8, wherein the channel estimator comprises a pilot detector (1222) for removing modulation on said received pilot signals and for obtaining an initial frequency response estimate, an inverse Fast Fourier Transform unit (1224) for performing an inverse Fast Fourier Transform on the initial frequency response estimate to obtain a channel impulse response estimate, a repetition processor (1226) for repeating the channel impulse response estimate, a combinerlfilter (1228) for performing at least one of combining or filtering the channel impulse response estimate to provide a full channel impulse response estimate, a threshold and padding processor (1230) for performing at least one of thresholding or zero-padding to obtain a vector, and an estimator Fast Fourier Transform unit (1232) for performing a Fast Fourier Transform on the vector to obtain a final frequency response estimate. 11. The communication signal demodulating apparatus as claimed in claim 8, wherein the channel estimator (1220) is configured to employ time filtering techniques to address delay spread effects. 12. The communication signal demodulating apparatus as claimed in claim 7, wherein the processor (1 170) is further configured to uniformly space the received pilot symbols of the second group from the received pilot symbols of the first group. 13. The communication signal demodulating apparatus as claimed in claim 12, wherein the processor (1 170) is further configured to stagger the at least two groups of received pilot symbols according to a two symbol staggering pattern. 14. The communication signal demodulating apparatus as claimed in claim 12, wherein the processor (1 170) is further configured to stagger the at least two groups of received pilot symbols according to an n-symbol staggering pattern. 15. The apparatus as claimed in claim 5, wherein said means for dividing further comprises means for uniformly spacing the received pilot symbols of the second group from the received pilot symbols of the first group. 16. The apparatus as claimed in claim 15, wherein said means for dividing Wher comprises means for staggering the at least two groups of received pilot symbols according to a two symbol staggering pattern. 17. The apparatus as claimed in claim 15, wherein said means for dividing further comprises means for staggering the at least two groups of received pilot symbols according to an n-symbol staggering pattern. 18. The apparatus as claimed in claim 6, wherein said processor is configured to further perform steps being characterized by uniformly spacing the received pilot symbols of the second group from the received pilot symbols of the fust group. 19. The apparatus as claimed in claim 18, wherein said processor is configured to further perform steps being characterized by staggering the at least two groups of received pilot symbols according to a two symbol staggering pattern. 20. The apparatus as claimed in claim 18, wherein said processor is c ~ ~ g u rteo d M e r perform steps being characterized by staggering the at least two groups of received pilot symbols according to an n-symbol staggering pattern.

Full Text

The present invention relates to a method of estimating a channel in a wireless communication system and an apparatus thereof.
[0002] The present invention relates generally to data communication, and more
specifically to time filtering for excess delay mitigation in orthogonal frequency division multiplexing (OFDM) systems.
II. Background
[0003] OFDM is a multi-carrier modulation technique that effectively partitions the
overall system bandwidth into multiple (NF) orthogonal subbands. These subbands are also referred to as tones, subcarriers, bins, and frequency channels. With OFDM, each subband is associated with a respective subcarrier that may be modulated with data. Up to NF modulation symbols may be transmitted on the NF subbands in each OFDM symbol period. Prior to transmission, these modulation symbols are transformed to the time-domain using an NF-point inverse fast Fourier transform (IFFT) to obtain a "transformed" symbol that contains NF chips.
[0004] OFDM can be used to combat frequency selective fading, which is
characterized by different channel gains at different frequencies of the overall system bandwidth. It is well known that frequency selective fading causes intersymbol interference (ISI), which is a phenomenon whereby each symbol in a received signal acts as distortion to one or more subsequent symbols in the received signal. The ISI distortion degrades performance by impacting the ability to correctly detect the received symbols. Frequency selective fading can be conveniently combated with OFDM by repeating a portion of each transformed symbol to form a corresponding OFDM symbol. The repeated portion is commonly referred to as a cyclic prefix.
[0005] The length of the cyclic prefix (i.e., the amount to repeat for each OFDM
symbol) is dependent on delay spread. The delay spread of a wireless channel is the time span or duration of an impulse response for the wireless channel. This delay spread is also the difference between the earliest and latest arriving signal instances (or multipaths) at a receiver for a signal transmitted via the wireless channel by a transmitter. The delay spread of an OFDM system is the maximum expected delay spread of the wireless channels for all transmitters and receivers in the system. To allow all receivers in the system to combat ISI, the cyclic prefix length should be equal to or longer than the maximum expected delay spread. However, since the cyclic prefix represents an overhead for each OFDM symbol, it is desirable to have the cyclic prefix length be as short as possible to minimize overhead. As a compromise, the cyclic prefix length is typically selected such that the cyclic prefix contains a significant portion of all multipath energies for most receivers in the system.
[0006] An OFDM system can withstand a delay spread that is smaller than or equal to
the cyclic prefix length. When this is the case, the NF subbands are orthogonal to one another. However, a given receiver in the system may observe excess delay spread, which is a delay spread that is greater than the cyclic prefix length. Excess delay spread can cause various deleterious effects, such as ISI and channel estimation errors, both of which can degrade system performance as described below. There is therefore a need in the art for techniques to mitigate the deleterious effects of excess delay spread in an OFDM system.
SUMMARY
[0007] Techniques for transmitting pilot and estimating the response of a wireless
channel with excess delay spread are described herein.
[0008] In an aspect, a method of estimating a channel in a wireless communication
system comprises obtaining at least two groups of received pilot symbols for at least two sets of pilot subbands, one group of received pilot symbols for each set of pilot subbands, wherein a second group is staggered with respect to a first group, dividing received pilot symbols into even symbols and odd symbols, wherein the even symbols include actual and excess components and the odd symbols include actual and excess components, determining an even effective estimate and an odd effective estimate based on the even symbols and the odd symbols, respectively, selecting a first set of time-filter coefficients for estimating an actual channel, selecting a second set of time-filter
coefficients for estimating an excess channel, time-filtering for an actual channel based on at least the even effective estimate, the odd effective estimate, the first set of time-filter coefficients and the second set of time-filter coefficients, and time-filtering for an excess channel based on at least the even effective estimate, the odd effective estimate, the first set of time-filter coefficients and the second set of time-filter coefficients.
[0009] In another aspect an apparatus in a wireless communication system comprising
means for obtaining at least two groups of received pilot symbols for at least two sets of pilot subbands, one group of received pilot symbols for each set of pilot subbands, wherein a second group is staggered with respect to a first group, means for dividing received pilot symbols into even symbols and odd symbols, wherein the even symbols include actual and excess components and the odd symbols include actual and excess components, means for determining an even effective estimate and an odd effective estimate based on the even symbols and the odd symbols, respectively, means for selecting a first set of time-filter coefficients for estimating an actual channel, means for selecting a second set of time-filter coefficients for estimating an excess channel, means for time-filtering for an actual channel based on at least the even effective estimate, the odd effective estimate, the first set of time-filter coefficients and the second set of time-filter coefficients, and means for time-filtering for an excess channel based on at least the even effective estimate, the odd effective estimate, the first set of time-filter coefficients and the second set of time-filter coefficients.
[0010] In yet another aspect, a computer readable media embodying a method for
estimating a channel in a wireless communication system comprising obtaining at least two groups of received pilot symbols for at least two sets of pilot subbands, one group of received pilot symbols for each set of pilot subbands, wherein a second group is staggered with respect to a first group, dividing received pilot symbols into even symbols and odd symbols, wherein the even symbols include actual and excess components and the odd symbols include actual and excess components, determining an even effective estimate and an odd effective estimate based on the even symbols and the odd symbols, respectively, selecting a first set of time-filter coefficients for estimating an actual channel, selecting a second set of time-filter coefficients for estimating an excess channel, time-filtering for an actual channel based on at least the even effective estimate, the odd effective estimate, the first set of time-filter coefficients and the second set of time-filter coefficients, and time-filtering for an excess channel based on
at least the even effective estimate, the odd effective estimate, the first set of time-filter
coefficients and the second set of time-filter coefficients.
[0011] Various aspects and embodiments of the invention are described in further
detail below.
BRIEF DESCRIPTION OF THE DRAWINGS
[0012] The features and nature of the present invention will become more apparent
from the detailed description set forth below when taken in conjunction with the
drawings in which like reference characters identify correspondingly throughout and
wherein:
[0013] ' FIG. 1 shows an OFDM modulator for an OFDM system;
[0014] FIGS. 2A and 2D show a wireless channel with excess delay spread and its
effective channel, respectively;
[0015] FIGS. 2B and 2C show a sequence of received chips for the wireless channel;
[0016] FIG. 3 shows a subband structure that may be used for the OFDM system;
[0017] FIG. 4 shows an access point and a terminal in the OFDM system; and
[0018] FIG. 5 shows a channel estimator.
FIG. 6 shows a terminal in a wireless communication system.
FIG. 7 shows a flowchart for a method of estimating a channel in a wireless communication system.
[0019] The word "exemplary" is used herein to mean "serving as an example,
instance, or illustration." Any embodiment or design described herein as "exemplary"
is not necessarily to be construed as preferred or advantageous over other embodiments
or designs.
[0020] The techniques described herein for time filtering for excess delay mitigation
may be used for various communication systems such as an orthogonal frequency
division multiplexing (OFDM)-based system, an Orthogonal Frequency Division
Multiple Access (OFDMA) system, a Code Division Multiple Access (CDMA) system,
a Time Division Multiple Access (TDMA) system, a Frequency Division Multiple
Access (FDMA) system, a single-input single-output (SISO) system, a multiple-input
multiple-output (MIMO) system, and so on.
[0021] In an OFDM system, a cyclic prefix is inserted at the beginning of each OFDM
symbol to remove interference across successive symbols. "When the delay spread of the channel is less than the cyclic prefix and the receiver is synchronized to choose the appropriate EFT window, there is no inter-symbol interference (ISI) between successive OFDM symbols. Further, linear convolution with the channel impulse response is
converted to a circular convolution,- and the orthogonality of the carriers is preserved.
In other words, there is no inter-carrier interference (ICI) between different carriers
within the same OFDM symbol.
[0022] When the delay spread of the channel exceeds the cyclic prefix, there is ICI as
well as ISI, and this could degrade the performance of the OFDM system. Increasing
the length of the cyclic prefix to avoid this degradation could lead to an unacceptable
overhead in the- system. .. In addition to the introduction of ICI/TSI, the presence of
excess delay spread could lead to further degradation in a coherent receiver that needs to
estimate the channel. Specifically, if the number of channel taps has increased and the
pilot resources allocated for channel estimation: could be insufficient. Clearly, the
degradation in such a scenario would depend on the allocated resources as well as the
amount of excess delay spread.
[0023] As with the cyclic prefix, increasing the resources for channel estimation may
lead to an unacceptable increase in overhead. Degradation in channel estimation could
be quite significant in some scenarios of practical interest, overshadowing the intrinsic
degradation due to ICI and ISI. Using channel estimation techniques that account for
the presence of excess delay spread mitigate such effects.
[0024] FIG. 1 shows a block diagram of an OFDM modulator 100 for an OFDM system.
The data to be transmitted is typically encoded and interleaved to generate code bits, which are then mapped to modulation symbols. The symbol mapping is performed by (1) grouping the code bits into B-bit binary values, where B ≥ 1, and (2) mapping each B-bit value to a specific, modulation symbol based on a modulation scheme (e.g., M-PSK or M-QAM, where M = 2B). Each modulation symbol is a complex value in a signal constellation corresponding to the modulation scheme. For each OFDM symbol period, one "transmit" symbol is sent on each of the NF subbands. Each transmit symbol can be either a modulation symbol for pilot/data or a signal value of zero (i.e., a "zero symbol"). An IFFT unit 110 performs an NFpornt IFFT on the NF transmit symbols for the NF total subbands in each OFDM symbol period and provides a transformed symbol that contains NF chips. The IFFT may be expressed as:
(Equation Removed)
where S is an NF x1 vector of transmit symbols for the NF subbands; WNFNF IS an NF xNF discrete Fourier transform (DFT) matrix;
s is an NF xl vector of time-domain chips; and "H " denotes the conjugate transpose.
The DFT matrix WNFXNF: is defined such that the (n,m) -th entry, wn,m, is given as:
(Equation Removed)
where n is a row index and m is a column index. WN xNp is an inverse DFT matrix.
[0025] A cyclic prefix generator 120 repeats a portion of each transformed symbol to obtain
a corresponding OFDM symbol that contains Nc chips, where Nc = NF + Ncp and Ncp
is the cyclic prefix length. An OFDM symbol period is the duration of one OFDM
symbol, which is Nc chip periods. The chips are conditioned and transmitted via a
wireless channel.
[0026] FIG. 2A shows an exemplary impulse response 210 of a wireless channel with
excess delay spread. Channel impulse response 210 includes two taps 212 and 214 for two multipaths in the wireless channel. Actual tap 212 has a complex gain of h1 and is located at tap index 1. Excess tap 214 has a complex gain of h, and is located at tap index Ne, which is outside of the cyclic prefix length Ncp. As used herein, "main channel" refers to the portion of the channel impulse response that is at or within the cyclic prefix length, "excess channel" refers to the portion of the channel impulse response that is outside of the cyclic prefix length, and "excess" refers to the difference between the tap index of an excess channel tap and the cyclic prefix length. For channel impulse response 210, the main channel includes one actual tap 212, the excess channel includes one excess tap 214, and the excess for tap 214 is Nex = Ne- Ncp.
[0027] FIG. 2B shows a sequence 220 of received chips for the wireless channel shown in
FIG. 2A. Received chip sequence 220 is a convolution of a transmitted chip sequence with taps 212 and 214 for the wireless channel. Received chip sequence 220 is composed of (1) a chip sequence 222 generated by convolving main channel tap 212 with the transmitted chip sequence and (2) a chip sequence 224 generated by convolving excess channel tap 214 with the transmitted chip sequence, where Si,- denotes the i-th chip for the current OFDM symbol, xi denotes the i-th chip for the previous OFDM symbol, and i = 1 .. Nc.
[0028] FIG. 2C shows the decomposition of received chip sequence 220 into different
components. Chip sequence 224 in FIG. 2B is replaced with (1) a chip sequence 226 generated by a circular convolution of excess channel tap 214 with the Nc chips for the current OFDM symbol, (2) a chip sequence 228 for the tail end of the previous OFDM symbol, and (3) a chip sequence 230 for the tail end of the current OFDM symbol. Chip sequences 222 and 226 represent the sequences that would have been received for taps 212 and 214 if the cyclic prefix length were sufficiently long and tap 214 is part of the main channel. However, since this is not the case, chip sequences 228 and 230 are both due to the excess delay spread. Chip sequence 228 represents the leakage of the previous OFDM symbol into the current OFDM symbol and is the source of intersymbol interference. Chip sequence 230 represents the disturbance to the circular convolution and is the source of intercarrier interference (ICI) and channel attenuation.
[0029] The intersymbol interference observed in each subband may be expressed as:
r
where X is an NF x1 vector of transmit symbols for the previous OFDM symbol; WH NexNF is an Nex xNF matrix with the last Nex rows of WHNFXNF ; and W1xNex (k) is a 1 x Nex vector with the first Nex elements of the k-th row of WNF xN.
The operation WHNEXNF X generates an Nex X1 vector XNex that contains the last Nex
chips of the previous OFDM symbol. The multiplication of XN with W1xNex (k)
generates the interference due to these last Nex chips on subband k.
[0030] The noise power on each subband due to intersymbol interference can be expressed
as:
(Equation Removed)
where Es is the transmit symbol energy, he2 is the power of the excess channel, and σ 2ISI, is the noise power due to ISI on each subband. As shown in equation (4), the ISI noise power per subband is (1) proportional to the excess channel energy he2, (2)
proportional to the excess Nex, which is indicative of the amount of leakage of the previous OFDM symbol onto the current OFDM symbol, and (3) inversely related to the
number of total subbands since the total ISI noise power is distributed over the NF
subbands.
[0031] The noise power on each subband due to intercarrier interference can be computed
in similar manner as for intersymbol interference and expressed as:
(Equation Removed)
where σ2ICI is the noise power due to ICI on each subband.
[0032] FIG. 2D shows an "effective" channel 240 for the wireless channel shown in FIG.
2A. Referring back to FIG. 2C, chip sequence 226 represents the contribution due to excess channel tap 214 (assuming that the cyclic prefix is long enough), and chip sequence 230 represents the source of ICI due to the excess channel. The subtraction operation for chip sequence 230 results partly in a reduction of the signal power for each subband. This subtraction can be accounted for by scaling down excess channel tap 214 by a factor of (1-Nex/NF). As shown in FIG. 2D, effective channel 240 includes tap 212 having the complex gain of h1 and a tap 216 having a complex gain of hc (1 - Nex /NF). The reduction in the gain of tap 216 relative to the gain of tap 214 is referred to as "channel attenuation" and results from excess delay spread for tap 214. The amount of attenuation is related to the excess Nex.
[0033] A receiver performs channel estimation in order to derive a channel estimate for the
wireless channel. Channel estimation is typically performed based on pilot symbols, which are modulation symbols that are known a priori by the receiver. The pilot symbols may be transmitted in various manners as described below.
[0034] FIG. 3 shows an exemplary subband structure that may be used for the OFDM
system. The OFDM system has an overall system bandwidth of BW MHz, which is partitioned into NF orthogonal subbands using OFDM. Each subband has a bandwidth of BW/NF MHz. For a spectrally shaped OFDM system, only NU of the NF total subbands are used for data/pilot transmission, where NU to allow the system to meet spectral mask requirements. For simplicity, the following description assumes that all NF subbands may be used in the OFDM system.
[0035] For the sake of illustration, an OFDM system is considered where channel
estimation is based on -uniformly spaced pilots in the frequency domain. The kth received OFDM symbol in the frequency domain can be written as
(Equation Removed)
where
• P is the number of pilots carriers, and D is the number of channel
taps assumed by the receiver.
• the vectors Y,H, w are of length P and the noise w is white complex
Gaussian with variance N0.
■ the matrix WP,D is the PxD submatrix of the unnorinalized DFT
matrix
(Equation Removed)
where N is the total number of snbcarriers.
[0036] The number of channel taps D ≤ P. However, in an embodiment a longer
channel estimate for dealing with scenarios where the channel has a delay spread larger than the cyclic prefix. To get a longer channel estimate, pilots are staggered across successive OFDM symbols, i.e., the pilot carrier indices are changed in successive OFDM symbols as described below.
Staggered pilots •
[0037] For simplicity, assuming a two symbol staggering pattern: if the uniformly
spaced pilot carriers are of the formin the even symbols, they would be
in the odd symbols. "With such staggering, we can get an estimate of up to a length 2P by using the pilot observations from two neighboring OFDM symbols. Specifically, assume a channel with 2P time domain taps. Then

[0038] For farther simplicity, set , so that the staggering is between phases
0 and. though the above expression can be carried through for any . We then have
(Equation Removed)
[0039] Similarly, for the odd symbols,

(Equation Removed)
[0040] Thus, the pilot observations in the even and odd symbols can be written as

(Equation Removed)
where and the superscripts "a" and "e" denote the "actual"'and
"excess" taps that correspond to ' and ' respectively.
Actual tap and excess tap are as discussed above and as
shown in, for example, FIG. 2A, where actual tap 212 and excess
tap 214 are shown. "Excess" refers to the difference between the
tap index of an excess channel tap and the cyclic prefix length. For
a channel impulse response, the main channel includes one actual
tap and the excess channel includes one excess tap, and the excess
for the excess tap is Nex = Ne - Ncp.
[0041] To get an estimate of the channel from the observations in Equation 7, one
possibility is to use a least-squares approach to estimate the effective time-domain channel. Equation 8 shows an even effective estimate and an odd effective estimate:
(Equation Removed)
[0042] The effective estimates above include both actual and excess components. A
simple way to get the full 2P tap channel estimate is
(Equation Removed)
[0043] Equation 9 is just a special case of a more general operation where the time-
domain estimates in Equation 8 (obtained every OFDM symbol) are averaged across multiple OFDM symbols. Such averaging is referred to as time-filtering, and it is done separately for each individual time-domain tap. The resulting estimate of tap l at any OFDM symbol m (odd or even) can be written as
(Equation Removed)
where Nf and Nb are the number of non-causal and causal taps, respectively. In this framework, Equation 9 corresponds to choosing Nf = 0, Nb = 2 and
(Equation Removed)
[0044] Thus, one set of time-filter coefficients is chosen for estimating the actual
channel (l

[0045] Consider more general strategies for choosing the time-filter coefficients for
the two halves. For clarity, the filter co-efficients for l

filter co-efficients for l ≥ P is denoted by ßn.
Time-filtering for the actual channel
[0046] Apart from separating the actual and excess channels, the choice of time-filter
coefficients are governed by other constraints as well. Time-filtering enables the capture of additional pilot energy and improvement in the reliability of channel estimates. However, using a long time-filter can introduce degradations due to time-variations of the channel.
[0047] For the sake of illustration, focus on the observed lth time-domain channel tap
in an even OFDM symbol, and assume that channel varies linearly over the Nf +Nb
symbols that are used by the time filter. Using Equation 8, we have
(Equation Removed)
where δa and δe are the slopes of the actual and excess channels at tap
I. Ideally, these time-variations would be canceled along with the excess channel. Hence, the constraints on the time-filter co-efficients can be summarized as:
(Equation Removed)
[0048] Since these constraints are invariant to a scale factor in the co-efficients, a
normalization constraint may be imposed, e.g. that the channel estimate be unbiased, which means
(Equation Removed)
[0049] For example, given a three tap filter with one non-causal tap,
i.s..Nf = 1,Nb =2, and the constraints in Equation 10 and Equation 11, the solution is
{0.25,0.5,0.25]. In the absence of excess channel taps, the optimal solution would be
{0.33,0.33,0.33}.
[0050] When the number of coefficients is greater than the number of linearly
independent constraints, the coefficients can be chosen to minimize the noise variance in the time-filtered estimate, i.e.,
(Equation Removed)
under the constraints of Equation 10 and Equation 11. It would be apparent to those skilled in the art that since the constraints are linear and the objective function is quadratic, this optimization can be solved using Lagrange multiplier techniques.
Time-filtering for the excess channel
[0051] Thus far, selection of filter taps has been restricted to the first P taps. For
l > P, the taps correspond to the excess channel, and are denoted by{ßn}.
[0052] In choosing{ßn}, the goal is reversed from that for l

excess taps are kept and contributions from the first P taps are eliminated. Hence, the constraints in Equation 10 are modified as:
(Equation Removed)
[0053] Only the first constraint has changed, and a scale factor constraint as in
Equation 11 can be imposed. For the three tap non-causal filter, the solution for {B-1,,ß0,ß1}is {-0.25,0.5,-0.25}. It would be apparent to those skilled in the art that
similar solutions can be obtained for other filter lengths (and other staggering patterns) as well.
Efficient generation of frequency domain estimate
[0054] In a modem implementation, the channel estimate in the frequency domain is
finally obtained on a per-interlace basis. That is, to reduce the number of computations
involved in the FFT operation to get the channel estimates in the frequency domain, a
P pt FFT is performed on the time domain channel estimate (after introducing a
suitable phase ramp), thereby resulting in the channel estimates for the interlace of
interest. With the estimation of the channel taps corresponding to the excess delay in
the channel, there are 2P taps for the channel estimate in the time domain. A channel
estimate for the required interlace can be obtained with a single P pt FFT operation. In
particular, let the IP channel taps in the time domain be represented by
h = [ha he] where ha and he are each P length vectors. Given the frequency estimate
for the P subcarriers {d = 0,1,2, P-1) corresponding to the interlace
m (m = 0,l,...7), then the frequency domain channel estimate for the d"' carrier in mth interlace is given by
(Equation Removed)
[0055] The extra P taps of the channel result in some trivial complex multiplications
(in four out of eight values of m) and additions. The phase ramp operation followed by the P pt EFT would have been performed irrespective of the number of channel taps being P. However, not truncating the channel to P taps, thereby allowing the extra P taps, requires additional memory for the buffering purposes.
[0056] Several assumptions and imposed limitations in the above discussion were
made for the sake of illustration. Specifically,
[0057] Staggering pattern: A simplistic staggering pattern with just two phases (0 and
4) was assumed. It would be apparent to those skilled in the art that the disclosed embodiments generalize to any other staggering pattern across different OFDM symbols. In each symbol, the pilots are uniformly spaced so that the excess channel aliases in the time-domain. The choice of the staggering pattern could be based on other factors and is of interest in itself.
[0058] Least-squares criterion: In going from the pilots in the frequency domain to the
aliased time domain channel estimate, a least-squares approach is used, which translates to an IFFT. It would be apparent to those skilled in the art that other criteria for deriving the time-domain estimate are possible, e.g. an MMSE approach.
[0059] A key point here is the relationship between the time-domain channel and pilot
observations that is induced by staggering. See Equation 7.
[0060] Time-filter length: A three tap filter for illustration was assumed. Clearly, the
approach is applicable for any number of taps that is greater than two, and the filter can total number of pilot observations is greater than the total channel length assumed, otherwise, perfect estimation of the complete channel is not possible.
[0061] Filter co-efficient selection: In choosing the filter co-efficients in accordance
with an embodiment, it is assumed that the same set of co-efficients are used for all the taps in the actual channel, and a different set is used for all the taps in the excess channel. In another embodiment, a different set of co-efficients is used for each tap in the actual channel as well as each tap in the excess channel (resulting in 2P sets of filter co-efficients in the example). Additional constraints have been imposed that the
lime-variation of the channel must be cancelled or suppressed when choosing the co- •
efficients. These constraints can be released depending on the number of time-filter co
efficients or other system design requirements.
[0062] Linear variation model: Finally, in formulating the constraints in Equation 10
etc, a model has been used where the channel varies linearly over the duration of
interest. Other approaches can be used to derive the constraints, e.g. a statistical model
can be assumed for the channel correlation over time and the problem can be posed in
terms of minimizing the variance of the time-variation errors.
[0063] For clarity, the pilot transmission and channel" estimation techniques have been
described for an OFDM system. These techniques may be used for other multi-carrier
modulation techniques such as discrete multi tone (DMT).
[0064] FIG. 4 shows a block diagram of an access point 1100 and a terminal 1150 in the
OFDM system. On the downlink, at access point 1100, a transmit (TX) data processor
1110 receives, formats, codes, interleaves, and modulates (i.e., symbol maps) traffic
data and provides modulation symbols (or simply, "data symbols"). An OFDM
modulator 1120 receives the data symbols and pilot symbols, performs OFDM
modulation as described for FIG. 1, and provides a stream of OFDM symbols. Pilot
symbols are transmitted in a staggered manner. A transmitter unit (TMTR) 1122
receives and converts the stream of OFDM symbols into one or more analog signals,
conditions (e.g., amplifies, filters, and frequency upconverts) the analog signals to
generate a downlink signal, and transmits the signal via an antenna 1124 to the
terminals.
[0065] At terminal 1150, an antenna 1152 receives the downlink'signal and provides a
received signal to a receiver unit (RCVR) 1154. Receiver unit 1154 conditions (e.g.,
filters, amplifies, and frequency downconverts) the received signal, digitizes the
conditioned signal, and provides received chips to an OFDM demodulator 1156.
[0066] FIG. 5 shows an embodiment of OFDM demodulator 1156. A cyclic prefix
removal unit 1212 removes the cyclic prefix appended to each OFDM symbol. An FFT unit 1214 then transforms each received transformed symbol to the frequency domain using an NF-point FFT and obtains NF received symbols for the NF subbands. FFT unit 1214 provides received pilot symbols to a processor 1170 and received data symbols to a detector 1216. Detector 1216 further receives a frequency response estimate Hm ,d(k)
for the downlink from processor 1170, performs detection on the received data symbols

WE CLAIM:
1. A method of estimating a channel in a wireless communication system in a
processor (1 130, 1170), wherein the estimation of a channel in a wireless
communication system is characterized by:
obtaining at least two groups of received pilot symbols for at least two sets of
pilot subbands, one group of received pilot symbols for each set of pilot subbands,
wherein the second group is staggered with respect to the first group;
dividing received pilot symbols into even symbols and odd symbols, wherein
the even symbols include actual and excess components and the odd symbols include
actual and excess components;
determining an even effective estimate and an odd effective estimate based on
the even symbols and the odd symbols, respectively;
selecting a first set of time-filter coefficients for estimating an actual channel;
selecting a second set of time-filter coefficients for estimating an excess
channel;
time-filtering said actual channel based on at least the even effective estimate,
the odd effective estimate, the first set of time-filter coefficients and the second set of
time-filter coefficients; and
time-filtering said excess channel based on at least the even effective estimate,
the odd effective estimate, the first set of time-filter coefficients and the second set of
time-filter coefficients,
wherein said actual component include data at or within a prefix length, and
excess components include data outside of the prefix length.
2. The method as claimed in claim 1, wherein said dividing further comprises
uniformly spacing the received pilot symbols of the second group from the received
pilot symbols of the first group.
3. The method as claimed in claim 2, wherein said dividing further comprises
staggering the at least two groups of received pilot symbols according to a two
symbol staggering pattern.
4. The method as claimed in claim 2, wherein said dividing M e r comprises
staggering the at least two groups of received pilot symbols according to an n-symbol
(n>) staggering pattern.
5. An apparatus in a wireless communication system, the apparatus characterized
by:
means for obtaining at least two groups of received pilot symbols for at least
two sets of pilot subbands, one group of received pilot symbols for each set of pilot
subbands, wherein the second group is staggered with respect to the first group;
means for dividing received pilot symbols into even symbols and odd
symbols, wherein the even symbols include actual and excess components and the
odd symbols include actual and excess components;
means for determining an even effective estimate and an odd effective
estimate based on the even symbols and the odd symbols, respectively;
means for selecting a first set of time-filter coefficients for estimating an
actual channel;
means for selecting a second set of time-filter coefficients for estimating an
excess channel;
means for time-filtering said actual channel based on at least the even
effective estimate, the odd effective estimate, the first set of time-filter coefficients
and the second set of time-filter coefficients; and
means for time-filtering said excess channel based on at least the even
effective estimate, the odd effective estimate, the first set of time-filter coefficients
and the second set of time-filter coefficients,
wherein said actual components include data at or within a prefix length, and
excess components include data outside of the prefix length.
6. An apparatus for estimating a channel in a wireless communication system,
the apparatus comprising:
a memory unit (1 132, 1 172);
a processor (1 130, 1 170) coupled to the memory unit (1 132, 1 172), wherein
the processor is configured to perform steps being characterized by:
obtaining at least two groups of received pilot symbols for at least two sets of
pilot subbands, one group of received pilot symbols for each set of pilot subbands,
wherein the second group is staggered with respect to the first group;
dividing said received pilot symbols into even symbols and odd symbols,
wherein the even symbols include actual and excess components and the odd symbols
include actual and excess components;
determining an even effective estimate and an odd effective estimate based on
the even symbols and the odd symbols, respectively;
selecting a first set of time-filter coefficients for estimating an actual channel;
selecting a second set of time-filter coefficients for estimating an excess
channel;
time-filtering said actual channel based on at least the even effective estimate,
the odd effective estimate, the first set of time-filter coefficients and the second set of
time-filter coefficients; and
time-filtering said excess channel based on at least the even effective estimate,
the odd effective estimate, the first set of time-filter coefficients and the second set of
time-filter coefficients,
wherein said actual components include data at or within a prefix length, and
excess components include data outside of the prefix length.
7. A communication signal demodulating apparatus comprising an OFDM
demodulator (1 156) configured to receive data comprising a plurality of symbols, the
plurality of symbols comprising pilot symbols and data symbols, the OFDM
demodulator (1 156) being characterized by:
a cyclic prefix removal unit (1 2 12) for removing a cyclic prefix appended to
said plurality of symbols;
a Fast Fourier Transform unit (1214) for transforming said plurality of
symbols into the frequency domain; and
a detector (1 2 16); and
a processor (1 170) configured to receive said pilot symbols from the Fast
Fourier Transform unit (1 2 14) and perform channel estimation on said received pilot
symbols by dividing said received pilot symbols into even symbols and odd symbols,
wherein the even symbols include actual and excess components and the odd symbols
include actual and excess components, and determining an even effective estimate
and an odd effective estimate based on the even symbols and the odd symbols,
respectively, the processor (1170) further configured to determine at least one
frequency response value based on the even effective estimate and the odd effective
estimate,
wherein the detector (1216) is configured to receive said data symbols from
the Fast Fourier Transform unit (1214) and the at least one frequency response value
from the processor (1 170) to obtain estimates of transmitted data symbols and provide
detected symbols, and
wherein said actual components include data at or within a prefix length, and
said excess components include data outside of the prefix length.
8. The communication signal demodulation apparatus as claimed in claim 7,
wherein the processor (1 170) comprises a channel estimator (1220) configured to
receive pilot symbols and perform channel estimation.
9. The communication signal demodulating apparatus as claimed in claim 8,
wherein the channel estimator (1220) comprises a pilot detector (1222) having an
output operably connected to an input of an inverse Fast Fourier Transform unit
(1224) having an output operably connected to an input of a repetition processor
(1226) having an output operably connected to an input of a combinerlfilter (1228)
having an output operably connected to an input of a threshold and padding processor
(1230) having an output operably connected to an input of an estimator Fast Fourier
Transform unit (1 232).
10. The communication signal demodulating apparatus as claimed in claim 8,
wherein the channel estimator comprises a pilot detector (1222) for removing
modulation on said received pilot signals and for obtaining an initial frequency
response estimate, an inverse Fast Fourier Transform unit (1224) for performing an
inverse Fast Fourier Transform on the initial frequency response estimate to obtain a
channel impulse response estimate, a repetition processor (1226) for repeating the
channel impulse response estimate, a combinerlfilter (1228) for performing at least
one of combining or filtering the channel impulse response estimate to provide a full
channel impulse response estimate, a threshold and padding processor (1230) for
performing at least one of thresholding or zero-padding to obtain a vector, and an
estimator Fast Fourier Transform unit (1232) for performing a Fast Fourier Transform
on the vector to obtain a final frequency response estimate.
11. The communication signal demodulating apparatus as claimed in claim 8,
wherein the channel estimator (1220) is configured to employ time filtering
techniques to address delay spread effects.
12. The communication signal demodulating apparatus as claimed in claim 7,
wherein the processor (1 170) is further configured to uniformly space the received
pilot symbols of the second group from the received pilot symbols of the first group.
13. The communication signal demodulating apparatus as claimed in claim 12,
wherein the processor (1 170) is further configured to stagger the at least two groups
of received pilot symbols according to a two symbol staggering pattern.
14. The communication signal demodulating apparatus as claimed in claim 12,
wherein the processor (1 170) is further configured to stagger the at least two groups
of received pilot symbols according to an n-symbol staggering pattern.
15. The apparatus as claimed in claim 5, wherein said means for dividing further
comprises means for uniformly spacing the received pilot symbols of the second
group from the received pilot symbols of the first group.
16. The apparatus as claimed in claim 15, wherein said means for dividing Wher
comprises means for staggering the at least two groups of received pilot symbols
according to a two symbol staggering pattern.
17. The apparatus as claimed in claim 15, wherein said means for dividing further
comprises means for staggering the at least two groups of received pilot symbols
according to an n-symbol staggering pattern.
18. The apparatus as claimed in claim 6, wherein said processor is configured to
further perform steps being characterized by uniformly spacing the received pilot
symbols of the second group from the received pilot symbols of the fust group.
19. The apparatus as claimed in claim 18, wherein said processor is configured to
further perform steps being characterized by staggering the at least two groups of
received pilot symbols according to a two symbol staggering pattern.
20. The apparatus as claimed in claim 18, wherein said processor is c ~ ~ g u rteo d
M e r perform steps being characterized by staggering the at least two groups of
received pilot symbols according to an n-symbol staggering pattern.

Documents:

4375-DELNP-2006-Abstract-(20-04-2010).pdf

4375-delnp-2006-abstract.pdf

4375-DELNP-2006-Claims-(16-09-2010).pdf

4375-DELNP-2006-Claims-(20-04-2010).pdf

4375-delnp-2006-claims.pdf

4375-DELNP-2006-Correspondence-Others-(05-05-2010).pdf

4375-DELNP-2006-Correspondence-Others-(16-09-2010).pdf

4375-DELNP-2006-Correspondence-Others-(20-04-2010).pdf

4375-delnp-2006-correspondence-others.pdf

4375-DELNP-2006-Description (Complete)-(20-04-2010).pdf

4375-delnp-2006-description (complete).pdf

4375-DELNP-2006-Drawings-(20-04-2010).pdf

4375-delnp-2006-drawings.pdf

4375-DELNP-2006-Form-1-(20-04-2010).pdf

4375-delnp-2006-form-1.pdf

4375-delnp-2006-form-18.pdf

4375-DELNP-2006-Form-2-(20-04-2010).pdf

4375-delnp-2006-form-2.pdf

4375-DELNP-2006-Form-3-(20-04-2010).pdf

4375-delnp-2006-form-3.pdf

4375-delnp-2006-form-5.pdf

4375-DELNP-2006-GPA-(20-04-2010).pdf

4375-delnp-2006-gpa.pdf

4375-delnp-2006-pct-210.pdf

4375-delnp-2006-pct-304.pdf

4375-DELNP-2006-Petition-137-(20-04-2010).pdf

4375-delnp-2006-petition-138.pdf

7375-DELNP-2006-Correspondence-Others-(26-04-2010).pdf

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Patent Number

257174

Indian Patent Application Number

4375/DELNP/2006

PG Journal Number

37/2013

Publication Date

13-Sep-2013

Grant Date

09-Sep-2013

Date of Filing

28-Jul-2006

Name of Patentee

QUALCOMM INCORPORATED

Applicant Address

5775 MOREHOUSE DRIVE, SAN DIEGO, CALIFORNIA 92121-1714, U.S.A.

Inventors:

#	Inventor's Name	Inventor's Address
1	FUYUN LING	17264 SANGALLO LANE, SAN DIEGO, CALIFORNIA 92127, U.S.A.
2	KIRAN MUKKAVILLI	7405 CHARMANT DRIVE, #2002, SAN DIEGO, CALIFORNIA 92122, U.S.A.
3	RAGHURAMAN KRISHNANMOORTHI	8730 COSTA VERDE BOULEVARD, #2302, SAN DIEGO, CALIFORNIA 92122, U.S.A.
4	DHANANJAY ASHOK GORE	8465 RAGENTS ROAD, #436, SAN DIEGO, CALIFORNIA 92122, U.S.A.
5	ASHOK MANTRAVADI	7855 AVENIDA, #229, SAN DIEGO, CALIFORNIA 92122, U.S.A

PCT International Classification Number

H04L 25/02

PCT International Application Number

PCT/US2005/001588

PCT International Filing date

2005-01-20

PCT Conventions:

#	PCT Application Number	Date of Convention	Priority Country
1	11/022,513	2004-12-22	U.S.A.
2	60/540,087	2004-01-28	U.S.A.