Title of Invention  A METHOD FOR ESTIMATING AN ENHANCED ESTIMATE OF A FREQUENCY RESPONSE OF A WIRELESS COMMUNICATION CHANNEL 

Abstract  ABSTRACT CHANNEL ESTIMATION FOR OFDM COMMUNICATION SYSTEMS The present invention relates to techniques to estimate the frequency response of a wireless channel in an OFDM system. In one method, an initial estimate of the frequency response of the wireless channel is obtained for a first group of subbands based on a pilot transmission received via the subbands in the first group. An estimate of the impulse response of the wireless channel is then derived based on the initial frequency response estimate. An enhanced estimate of the frequency response of the wireless channel is then derived for a second group of subbands based on the impulse response estimate. The first and second groups may each include all or only a subset of the usable subbands. Subband multiplexing may be used to allow simultaneous pilot transmissions by multiple terminals on their associated groups of subbands. Fig. 5 
Full Text  RELATED APPLICATIONS This application is related to both U.S. Provisional Patent Application Serial No. 60/422,362, filed October 29, 2002, entitled "Channel Estimation For OFDM Communication Systems," and to U.S. Provisional Patent Application Serial No. 60/422,368, filed October 29, 2002, entitled "Uplink Pilot And Signaling Transmission In Wireless Communication Systems," filed on October 29, 2002, which arc ■ incorporated herein by reference in its entirety for all purposes. BACKGROUND I. Field of the Invention The present invention relates generally to data communication, and more specifically to techniques for estimating the response of a wireless channel in a communication system with multiple subbands, such as an orthogonal frequency division multiplexing (OFDM) system. II. Background Wireless communication systems are widely deployed to provide various types of communication such as voice, packet data, and so on. These systems may be multipleaccess systems capable of supporting communication with multiple users by sharing the available system resources. Examples of such multipleaccess systems include code division multiple access (CDMA) systems, time division multiple access (TDMA) systems, and orthogonal frequency division multiple access (OFDMA) systems. OFDM effectively partitions the overall system bandwidth into a number of (N) orthogonal subbands. These subbands are also referred to as tones, frequency bins, and frequency subchannels. With OFDM, each subband is associated with a respective subcarrier upon which data may be modulated. Each subband may thus be viewed as an independent transmission channel that may be used to transmit data. 5] In a wireless communication system, an RF modulated signal from a transmitter may reach a receiver via a number of propagation paths. For an OFDM system, the N ubbands may experience different effective channels due to different effects of fading and muitipath and may consequently be associated with different complex channel pains. An accurate estimate of theresponse of the wireless channel betv/een the ransmitter and the receiver is normally needed in order to effectively transmit data on he available subbands. Channel estimation is typically performed by sending a piloi Tom the transmitter and measuring the pilot at the receiver. Since the pilot is made up of symbols that are known a priori by the receiver, the channel response con be estimated as the ratio of the received pilot symbol over the transmitted pilot symbol for each subband used for pilot transmission. Pilot transmission represents overhead in the OFDM system. Thus, it is desirable to minimize pilot transmission to the extent possible. However, because of noise and other artifacts in the wireless channel, a sufficient amount of pilot needs to be transmitted in order for the receiver to obtain a reasonably accurate estmiate of \\\c. channel response. Moreover, the pilot transmissions need to be repeated to account for variations in the channel over time due to fading and changes in the muitipath constituents. Consequently, channel estimation for an OFDM system normally consumes a noticeable portion of the system resources. In the downlink of a wireless communication system, a single pilot transmission from an access point (or a base station) can be used by a number of terminals in estimate the response of the distinct downlink channels from the access point to each of the terminals. However, in the uplink, each tenninal needs to send a pilol (ransmission separately in order to enable the access point to estimate the uplink chnnnel from the terminal to the access point. Consequently, the overhead due to pilol transmissions is exacerbated due to uplink pilot transmissions. There is therefore a need in the art for techniques to more efficiently estimate the channel response in an OFDM system, particularly in the uplink. SUMMARY Techniques are provided herein to estimate the frequency response of a wireless channel in a communication system with multiple subbands (e.g., an OH)M system). It is recognized that the impulse response of the wireless channel can be characterized by L taps, where L is typically much less than the N total subbands in the OFDM system. Because only L taps is needed for the channel impulse response, the frequency resp(iM.se of the wireless channel lies in a subspace of dimension L (instead of N) and may be fully characterized based on the channel gains for as few as L appropriately selected subbands (instead of all N subbands). Moreover, even when more than I. channel I'aie^s are available, the property described above may be used to obtain an enhanced estiiriale of the frequency response of the wireless channel by suppressing the noise components outside this subspace, as described below. In one embodiment, a method is provided for estimating the frequency response of the wireless channel (e.g., in the OFDM system). In accordance with the method, nn initial estimate of the frequency response of the wireless channel is obtained for a first group of subbands based on a pilot transmission received via the subbands in the first group. The first group may include all or only a subset of the subbands usab]e for data transmission. An estimate of the impulse response of the wireless channel us then derived based on the initial frequency response estimate and a first discrete Fourier transform (DFT) matrix for the subbands in the first group. The impulse response estimate may be derived as a least square estimate, as described below. An enhanced estimate of the frequency response of the wireless channel is then derived for a second group of subbands based on the impulse response estimate and a second DFI^ matrix for the subbands in the second group. The second group may include all or a subset of the usable subbands, and would include at least one additional subband not included in the first group if this first group does not include all usable subbands. Various aspects and embodiments of the invention are described in further detail below BRIEF DESCRIPTION OF THE DRA WlNGvS The features, nature, and advantages of the present invention will bec(^me more apparent from the detailed descripdon set forth below when taken in eonjnnclion with the drawings in which like reference characters identify correspondingly throughout and wherein: FIG. 1 shows an OFDM subband structure; FIG. 2A shows the relationship between the frequency response and the impulse response of a wireless channel; FIG. 2B shows a DFT matrix for the N total subbands in the OFDM system; FIG. 3A shows the relationship between the DFT matrices for the M usable ibbands and the N total subbands in the OFDM system; FIG. 3B shows derivation of an enhanced frequency response estimate based on n impulse response estimate derived from pilot transmission on the M usable subhancis; FIG. 4A shows the relationship between the DFT matrices for vS assigned ubbands and the N total subbands; FIG. 4B shows derivation of the enhanced frequency response estimate based on m impulse response estimate derived from pilot transmission on the S assigned ubband; FIG. 5 shows an OFDM subband structure that supports subband multiplexing; FIG. 6 shows a process for estimating the frequency response of the wireless channel; and FIG. 7 shows a block diagram of an access point and a terminal. DETAILED DESCRIPTION The channel estimation techniques described herein may be used for any communication system with multiple subbands. For clarity, these techniques are described for an OFDM system. FIG. 1 shows a subband structure'lOO that may be used for an OFDM system. The OFDM system has an overall system bandwidth of W MIlz, which is partitioned into N orthogonal subbands using OFDM. Each subband has a bandwidth nf W/N MHz. In a typical OFDM system, only M of the N total subbands arc used for data transmission, where M example, if the system bandwidth is W = 20 MHz and N = 256, then the bandwidth of each subband is 78.125 KHz (or W/N MHz) and the duration of each transformed symbol is 12.8 ixsec (or N/W jxsec). OFDM can provide certain advantages, such as the ability to combat frequciicy selective fading, which is characterized by different channel gmns at different Frequencies of the overall system bandwidth. It is well known that frequency selective fading is accompanied by intersymbol interference (ISI), which is a phenmnenun whereby each symbol in a received signal acts as distortion to subsequent symbols in the received signal. The ISI distortion degrades performance by impacting the ability to correctly detect the received symbols. Frequency selective fading can be conveniently combated with OFDM by repeating a portion of (or appending a cyclic prefix to) each transformed symbol to fonn a corresponding OFDM symbol, which is then transmitted over a wireless channel. The length of the cyclic prefix (i.e., the amount to repeat) for each OFDfvl symbol is dependent on the delay spread of the system. The delay spread for a given transmitter is the difference between the earliest and latest arriving signal instances at Each transformed symbol has a duration of N sample periods, where eacii sample period has a duration of (1/W) sec, The cyclic prefix may be defined to include Cp samples, where Cp is a suitable integer selected based on the delay spread ot the system. In particular, Cp is selected to be greater than or equal to the number of taps (L) for the impulse response of the wireless channel (i.e., Cp > L). In this case, each OFDM symbol would include N+ Cp samples, and each symbol period would span N + Cp sample periods. The N subbands of the OFDM system may experience different channel conditions (i.e., different effects due to fading and multipath) and may be associated with different complex channel gains. An accurate estimate of the channel response is normally needed in order to property process (e.g., decode and demodulate) data at the receiver. The wireless channel in the OFDM system may be characterized by either a timedomain channel impulse response, h, or a corresponding frequencydomain channel frequency response, H. The channel frequency response H is the discrete Fourier transform (DFT) of the channel impulse response h . This relationship may be expressed in matrix form, as follows: he The vector h includes one nonzero entry for each tap of the channel impulse response. Thus, if the channel impulse response includes L taps, where T. FIG. 2A graphically shows the relationship between the channel frequency response H and the channel impulse response h. The vector h includes N timedomain values for the impulse response of the wireless channel from the transmitter to the receiver. This vector h can be transformed to the frequency domain by pre multiplying it with the DFT matrix W. The vector H includes N frequencydonuni! values for the complex channel gains of the N subbands. FIG. 2B graphically shows the matrix W, which is an (NxN) tnalrix comprised of the elements defined in equation (2). Techniques are provided herein to obtain an enhanced estimate of the frequeary response of the wireless channel in the OFDM.system. It is recognized that the impalse. response of the wireless channel can be characterized by L taps, where L is typically much less than the number of total subbands in the system (i.e., L Because only L taps are needed for the channel inripulse response, the channel frequency response H lies in a subspace of dimension L (instead of N). More specifically, the frequency response of the wireless channel may be fully characterized based on the channel gains for as few as L appropriately selected subbands, instend of all N subbands. Even if more than L channel gains are available, an enhanced estimate of the frequency response of the wireless channel may be obtained by suppressing the noise components outside this subspace, as described below. The model for the OFDM system may be expressed as; where r is a "receive" vector with N entries for the symbols received on the N subbands; X is a ^'transmit" vector with N entries for the symbols transmitted on the N subbands (the entries for the unused subbands are zeros); n is a vector with entries for additive white Gaussian noise (AWGN) received on the N subbands; and "o" denotes the Hadmard product (i.e., a pointwise product, where the /th element of r is the product of the ilh elements of x and H). The noise n is assumed to have zero mean and a variance of cr'. The channel estimation techniques described herein may be used m conjunction with various pilot transmission schemes. For clarity, these techniques me described for two specific pilot transmission schemes. An enhanced estimate of the frequency response of the wireless channel, /\ may then be derived from the least square channel impulse response estimate, h follows: where Hl is an (Mxl) vector for the enhanced channel frequency response estimate. Equation (7) indicates that the enhancec bannel frequency response estimate H^/ may be obtained for all M data subbands basedof the least square channel impulse response estimate h that includes only L entries, where L FIG. 3B graphically shows the relationship between the. enhanced channel frequency response estimate H^ and the least square channel impulse response estiniatc ^ Is ^ Is h^ . The vector h^ includes L timedomain values for the least squ^u^e channel impulse response estimate. This vector h^ can be transformed to the frequencydomain by i)rc multiplying it with the matrix W. The resultant vector H, includes M iVequency doraain values for the complex gains for the M data subbands. For clarity, the channel estimation techniques are described above with three distinct steps: 1. Obtain the initial channel frequency response estimate H^^: .^ b 2. Derive the least square channel impulse response estimate h^^ based on the initial channel frequency response estimate H^ ; and ' V ■ ' ^ Is 3. Derive the enhanced channel frequency response estimate H^^ based on the channel impulse respoffseestimate.h^ . The channel estimation may also be j)erformed such that a step may be implicidv (instead of explicitly) performed. In particular, the enhanced channel freque^ncy ^ Is response estimate H^ may be derived directly from the initial channel frequency response estimate H^, as follows: where P^ is the transmit power used for the pilot symbol in each of the M data subbands. It can be shown that the MSE in equadon (9) is the trace of the noise covariance matrix after the least square operadon (i.e., the covariance matrix of . In a second pilot transmission scheme, pilot symbols are transmitted on each of S designated subbands, where S L. Typically, the number of designated subbands is less than the number of data subbands (i.e., S downlink, the other (M—S) data subbands may be used to transmit traffic data and/or overhead data. On the uplink, the M data subbands may be partitioned into disioint groups of S subbands, and each group may then be assigned to a different (enninal for pilot transmission. This subband multiplexing, whereby multiple tenninals transmit concurrendy on disjoint groups of subbands, may be used to improve system efficiency. For clarity, channel estimadon is described below for subband multiplexing whereby each designated terminal transmits a pilot only on its S assigned subbands. FIG. 4A graphically shows the relationship between the matrices W, and VV . The S rows of the matrix W,. are the S rows of the matrix W corresponding to the S subbands assigned to terminal i (which are shown as the unshaded rows). The L columns of the matrix W are the first L columns of the matrix W. vSince each terminal is assigned a different group of subbands for pilot transmission on the uplink. the matrix Wn is different for different terminals. Again, the optiraizadon in equation (11) is over all possible channel impulse responses hn. The least square channel impulse response estimate h for tenninal / is equal to the hypothesized response hf that results in the minimum error between Ihe initial frequency response estimate H,. and the frequency response corresponding to h ,, which is given by W_,hy. The solution to equation (11) may be expressed as: As shown in equadon (12), the least square channel impulse response estimate h, For terminal / may be derived based on the initial channel frequency response estimate H,, which is obtained based on the uplink pilot received on only the S subbands assigned to terminal i. In particular, the estimate h, may be obtained by performing a least square operation (i.e., a premultiplicadon with (W. W,)1W. ) on the initial estimate IT. The vector h, includes L entries for the L taps of the channel impulse response, where L estimate h^ , as follows: vhere H is an (Mxl) vector for the enhanced channel frequency response estimate for terminal i.  Is . 3quation (13) indicates that the enhanced channel frequency response estimate H, for :erminal i may be obtained for all M data subbands based on the least square channel impulse response estimate h, that includes only L entries, where typically L frequency response estimate H and the least square channel impulse response estimate ^h '^ is h, for terminal i. The vector h,. includes L timedomain values for the least square channel impulse response estimate for terminal i, This vector h, can be transformed to   h the frequency domain by premultiplying it with the DFT matrix W. The vector H, includes M frequencydomain values for the complex gains for the M data subbands for terminal i.  Is The enhanced channel frequency response estimate H may be derived directly from the initial channel frequency response estimate H, as follows: Equation (14) combines equations (12) and (13), and the derivation of the least square channel impulse response estimate h,. is implicitly performed. The quality of the enhanced estimate H is dependent on various factors, one oi which is whether all or only a subset of the N total subbands is used for data transmission. Each of these two cases is analyzed separately below. e distributed to the Q groups in some other manners, and this is within the scope of the ivention. The Q groups of subbands may he assigned to up to Q terminals for uphnk pilot ransmission. Each terminal would then transmit a pilot only on its S assigned ubbands. With subband multiplexing, up to Q terminals may simultaneously transmit )ilots on the uphnk on up to M usable subbands. This can greatly reduce the amount of overhead needed for uphnk pilot transmission. To allow the access point to obtain high quahty channel estimates, each terminal may increase the transmit power per subband by a factor of Q. This would result in the total energy for the pilot transmission on the S assigned subbands to be the same as if all M data subbands were used for pilot transmission. The same total pilot energy would allow the access point to estimate the channel response for all M usable subbands based on pilot transmission on only a subset of these subbands with little or no loss in quality, as described above. If subband multiplexing is used to permit simultaneous pilot transmission by multiple terminals, then the signals from nearby terminals may cause, substantial interference to the signals from faraway terminals if all terminals transmit at full power. In particular, it can be shown that frequency offset among the terminals can result in intersubband interference. This interference can cause degradation in the channel estimate derived from uplink pilots and/or increase the bit error rate of uplink data transmissions. To mitigate the effects of intersubband interference, the terminals may be power controlled so that the nearby terminals do not cause excessive interference to faraway terminals. The effect of interference from nearby terminals was investigated, and it was found that power control may be applied coarsely to mitigate intersubbanci interference. In particular, it was found that if the maximum frequency offset among the temiinals is 300 Hz or less in the case of the exemplary system with 256 total subbands in a 20 MHz channel, and Q = 12, then by hmiting the received signaltonoise ratios (SNRs) of the nearby terminals to 40 dB or less, there would be a loss of 1 dB or less in the .SNR:; of the other terminals. If the frequency offset among the terminals is 1000 M/. or less, tlirn the SNRs of the nearby tenninals should be limited to 27 dB to ensure [ dB or less of loss in the SNRs of the other terminals. If the SNR needed to achieve the highest rate supported by an OFDM system is less than 27 dB (40dB), then limiting the SNR of each perminal to 27 dB or less (or 40 dB or less) would not have any impact on the maximum upported rate for the nearby terminals. The coarse power control requirements stated above may be achieved wiil\ a low power control loop. For example, control messages may be sent when aiul as leeded to adjust the uplink power of nearby terminals (e.g., when the power level ;hanges due to movement by these terminals). Each terminal may be infonned of (lie nitial transmit power level to use for the uplink as part of a call setup procedure when accessing the system. The groups of subbands may also be assigned to the terminals in a manner to mitigate the effect of intersubband interference. In particular, terminals with high received SNRs may be assigned subbands that are near each other. Terminals with low received SNRs may be assigned subbands that are also near each other, but away from the subbands assigned to the terminals with high received SNRs. Certain benefits may be obtained from the subband grouping and uniform subband spacing described above. However, other channel grouping and spacing schemes may also be used, and this is within the scope of the invention. In general, the groups may include the same or different number of subbands, and the subbands in each group may be uniformly or nonuniformly distributed across the M usable subbands. FIG 6 is a flow diagram of an embodiment of a process 600 for estimating the frequency response of a wireless channel. Process 600 provides an enhanced channel frequency response estimate for all M data subbands based on pilot transmission received on S assigned subbands, where S An initial estimate of the frequency response of the wireless channel, H ■ , is first obtained for the S assigned subbands based on the pilot received on these S subbands, as shown in equation (10) (step 612). The DFT matrix W, is then formed and includes the first L columns of the matrix W and the S rows of the matrix W corresponding to the S subbands used for pilot transmission (step 614). A least square estimate of the impulse response of the wireless channel, h,. , is en derived based on the initial channel frequency response estimate H,. and the matrix V, as shown in equation (12) (step 616). The DFT matrix W is next formed and icludes the first L columns of the matrix W and the M rows of the matri;^ W orresponding to the M data subbands (step 618). In general, the matrix W can include ny combination of rows for any group of subbands for which the frequency response is esired. An enhanced estimate of the frequency response of the wireless channel, H. , is hen derived based on the least square channel impulse response estimate h, and the natrix W , as shown in equation (13) (step 620). The vector H. includes the complex ^ains for all subbands covered by the mai^^ W . The derivations for steps 616 and 620 nay be combined, as described above and.shown in equation (14). FIG. 7 is a block diagram of an eJnbodiment of an access point 700 and a terminal 750, which are capable of performing channel estimation described herein. On the downlink, at access point 700, traffic data is provided to a TX data processor 710, which formats, codes, and interleaves the traffic data to provide coded data. An OFDM modulator 720 then receives and processes the coded data and pilot symbols to provide a stream of OFDM symbols. The processing by OFDM moduialoE720 may include (1) symbol mapping the coded data to form modulation symbols, (21 multiplexing the modulation symbols with pilot symbols, (3) transforming the modulation symbols and pilot symbols to obtain transformed symbols, and (4) appending a cyclic prefix to each transformed symbol to form a corresponding OFDM symbol. For the downlink, the pilot symbols may be multiplexed with the modulalion symbols using, for example, time division multiplexing (TDM). For TDM, the pilot and modulation symbols are transmitted on different time slots. The pilot symbols may bt: transmitted on all M usable subbands or a subset of these subbands. A transmitter unit (TMTR) 722 then receives and converts the stream of OFDM symbols into one or more analog signals and further conditions (e.g., amplifies, filters, and frequency upconverts) the analog sigiA to generate a downlink modulated signal 44 to provide demodulated data, which are further processed by an RX data processor '46 to recover the transmitted traffic data. OFDM demodulator 744 may determine the nitial channel frequency response estimate H, for each designated terminal or provide he received pilot symbols that may be used; to obtain H, . A controller 730 receives H, (or equivalent information), detennines the least square channel impulse response h,. for designated active terminal based on H,, and further obtains the enhanced ^ Is '^Is ^ Is channel frequency response estimate H. based on h, . The enhanced estimate H, may thereafter be used for downlink data transmission to the terminal. Controllers 730 and 770 direct the operation at the access point and terminal, respectively. Memory units 732 and 772 provide storage for program codes and data used by controllers 730 and 770, respectively. The channel estimation techniques described herein may be implemenlccl by various means. For example, these techniques may be implemented \n hardware, software, or a combination thereof. For a hardware implementation, the elements used to implement any one or a combination of the techniques may be implemented within one or more application specific integrated circuits (ASICs), digital signal processors (DSPs), digital signal processing devices (DSPDs), programmable logic devices (PUDs), field programmable gate arrays (FPGAs), processors, controllers, micro^ controllers, microprocessors, other electronic units designed to perform the functions described herein, or a combination thereof. For a software implementation, the channel estimation techniques may be implemented with modules (e.g., procedures, functions, and so on) that perform the functions described herein. The software codes may be stored in a memory unit (e.g., memory units 732 or 772 in FIG. 7) and executed by a processor (e.g., controller 730 tn 770). The memory unit may be implemented within the processor or external to tlie processor, in which case it can be comimunicatively coupled to the processor via various means as is known in the art. 1 The previous description of the disclosed embodiments is provided to enable any person skilled in the art to make or use the present invention. Various modifications to these embodiments will be readily apparent to those skilled in the art, and the generic principles defined herein may be applied to other embodiments without departing from the spirit or scope of the invention. Thus, the present invention is not intended lo he limited to the embodiments shown herein but is to be accorded the widest scope consistent with the principles and novel features disclosed herein. WE CLAIM : 1. A method for estimating a frequency response of a wireless communication channel, the method comprising: estimating the frequency response of the wireless communication channel based on a subset of a plurality of available tones, wherein the subset comprises a plurality of equally spaced tones, the equally spaced tones being equally spaced by a spacing number of the plurality of available tones between the equally spaced tones. 2.The method as claimed in claim 1, wherein the estimating comprises: using a received power equal to P+N/S, where P is an average received power for the plurality of available tones, N is a total available tone number and S is a subset number indicating the subset number of tones in the subset. 3. The method as claimed in claim 1, wherein the subset comprises a subset number S of the plurality of available tones, wherein S = 2, wherein r is an integer. 4. The method as claimed in claim 1, wherein the subset consists of the plurality of equally spaced tones. 5. A wireless communication access point comprising: a receiver configured to receive a wireless communication signal in a wireless communication channel; and a processor coupled to the receiver and configured to estimate a frequency response of the wireless communication channel based on a subset of a plurality of available tones, wherein the subset comprises a plurality of equally spaced tones, the equally spaced tones being equally spaced by a spacing number of the plurality of available tones between the equally spaced tones. 6. The wireless communication access point of claim 5, wherein the processor is configured to use a received power equal to P*N/S for estimating the frequency response, where P is an average received power for the plurality of available tones, N is a total available tone number and S is a subset number indicating the subset number of tones in the subset. 7. The wireless communication access point of claim 5, wherein the subset comprises a subset number S of the plurality of available tones, wherein S = 2\ wherein r is an integer. 8. The wireless communication access point of claim 5, wherein the subset consists of the plurality of equally spaced tones. 9. Awireless communication access point comprising: a receiving means for receiving a wireless communication signal in a wireless communication channel; and a processing means for estimating a frequency response of the wireless communication channel based on a subset of a plurality of available tones, wherein the subset comprises a plurality of equally spaced tones, the equally spaced tones being equally spaced by a spacing number of the plurality of available tones between the equally spaced tones, the processing means coupled to the receiving means. 10. The wireless communication access point of claim 9, wherein the processing means is further configured to use a received power equal to P*N/S, where P is an average received power for the plurality of available tones, N is a total available tone number and S is a subset number indicating the subset number of tones in the subset for estimating the frequency response. 11. The wireless communication access point of claim 9, wherein the subset comprises a subset number S of the plurality of available tones, wherein S = 2\ wherein r is an integer. 12. The wireless communication access point of claim 9, wherein the subset consists of the plurality of equally spaced tones. 13. A computer readable medium embodying instructions for performing a method for estimating a frequency response of a wireless communication channel, the method comprising: spaced tones, the equally spaced tones being equally spaced by a spacing number of the plurality of available tones between the equally spaced tones. 14. The computer readable medium as claimed in claim 13, wherein the estimating comprises: using a received power equal to P*N/S, where P is an average received power for the plurality of available tones, N is a total available tone number and S is a subset number indicating the subset number of tones in the subset. 15. The computer readable medium as claimed in claim 13, wherein the subset comprises a subset number S of the plurality of available tones, wherein S = 2', wherein r is an integer. 16. The computer readable medium as claimed in claim 13, wherein the subset consists of the plurality of equally spaced tones. 17. A method of selecting a pilot signal for a pilot transmission from a wireless communication terminal, the method comprising: selecting for the pilot transmission a subset of a plurality of available tones, wherein the subset comprises a plurality of equally spaced tones, the equally spaced tones being equally spaced by a spacing number of the plurality of available tones between the equally spaced tones. 18. The method as claimed in claim 17, comprising: selecting a transmit power P equal to TN/S, where T is an average power for the plurality of available tones, N is a total available tone number and S is a subset number indicating the subset number of tones in the subset. 19. The method as claimed in claim 17, wherein the subset comprises a subset number S of the plurality of available tones, wherein S = 2', wherein r is an integer. 20. The method as claimed in claim 17, wherein the subset consists of the plurality of equally spaced tones. 21. A wireless communication terminal comprising: a processor configured to select for a pilot transmission a subset of a plurality of available tones, wherein the subset comprises a plurality of equally spaced tones, the equally spaced tones being equally spaced by a spacing number of the plurality of available tones between the equally spaced tones; and a transmitter coupled to the processor, the transmitter configured to transmit the pilot transmission over the air. 22. The wireless communication terminal as claimed in claim 21, wherein the processor is further configured to select a transmit power P equal to T*N/S, where T is an average power for the plurality of available tones, N is a total available tone number and S is a subset number indicating the subset number of tones in the subset. 23. The wireless communication terminal as claimed in claim 21, wherein the subset comprises a subset number S of the plurality of available tones, wherein S = 2\ wherein r is an integer. 24. The wireless communication terminal as claimed in claim 21, wherein the subset consists of the plurality of equally spaced tones. 25. A wireless communication terminal comprising: a processing means for selecting for a pilot transmission a subset of a plurality of available tones, wherein the subset comprises a plurality of equally spaced tones, the equally spaced tones being equally spaced by a spacing number of the plurality of available tones between the equally spaced tones; and a transmitting means for transmitting the pilot transmission over the air, the transmitting means coupled to the processing means. 26. The wireless communication terminal as claimed in claim 25, wherein the processing means is further configured to select a transmit power P equal to T+N/S, where T is an average power for the plurality of available tones, N is a total available tone number and S is a subset number indicating the subset number of tones in the subset. 27. The wireless communication terminal as claimed in claim 25, wherein the subset comprises a subset number S of the plurality of available tones, wherein S = 2', wherein r is an integer. 28. The wireless communication terminal as claimed in claim 25, wherein the subset consists of the plurality of equally spaced tones. 29. A computer readable medium embodying instructions for performing a method for selecting a pilot signal for a pilot transmission from a wireless communication terminal, the method comprising; selecting for the pilot transmission a subset of a plurality of available tones, wherein the subset comprises a plurality of equally spaced tones, the equally spaced tones being equally spaced by a spacing number of the plurality of available tones between the equally spaced tones. 30. The computer readable medium as claimed in claim 29, the method further comprising: selecting a transmit power P equal to TN/S, where T is an average power for the plurality of available tones, N is a total available tone number and S is a subset number indicating the subset number of tones in the subset. 31. The computer readable medium as claimed in claim 29, wherein the subset comprises a subset number S of the plurality of available tones, wherein S = 2\ wherein r is an integer. 32. The computer readable medium as claimed in claim 29, wherein the subset consists of the plurality of equally spaced tones. 

1867CHENP2007 AMENDED PAGES OF SPECIFICATION 09102012.pdf
1867CHENP2007 AMENDED CLAIMS 09102012.pdf
1867CHENP2007 AMENDED PAGES OF SPECIFICATION 18102012.pdf
1867CHENP2007 AMENDED CLAIMS 18102012.pdf
1867CHENP2007 EXAMINATION REPORT REPLY RECEIVED 09102012.pdf
1867CHENP2007 FORM1 18102012.pdf
1867CHENP2007 FORM3 09102012.pdf
1867CHENP2007 OTHER PATENT DOCUMENT 09102012.pdf
1867CHENP2007 CORRESPONDENCE OTHERS 09042012.pdf
1867CHENP2007 EXAMINATION REPORT REPLY RECEIVED 18102012.pdf
1867chenp2007assignement.pdf
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1867chenp2007description(complete).pdf
Patent Number  254343  

Indian Patent Application Number  1867/CHENP/2007  
PG Journal Number  44/2012  
Publication Date  02Nov2012  
Grant Date  26Oct2012  
Date of Filing  01May2007  
Name of Patentee  QUALCOMM INCORPORATED  
Applicant Address  5775 MOREHOUSE DRIVE, SAN DIEGO, CALIFORNIA 92121, USA  
Inventors:


PCT International Classification Number  H04B 7/05  
PCT International Application Number  PCT/US03/34506  
PCT International Filing date  20031029  
PCT Conventions:
