Title of Invention

A METHOD OF PROVIDING FEEDBACK IN RADIO FREQUENCY RECEIVER AND RADIO FREQUENCY RECEIVER THEREOF

Abstract A method and system for increasing the compression point of a receiver by deriving a feedback signal from mixer output signals. The feedback signal prevents the receiver from going into compression on strong out-of-band or blocking signals, while enhancing the receiver gain at the desired frequency. The desired frequency coincides with the local oscillator (LO) signal and is therefore particularly applicable for, but not limited to, homodyne receivers where selectivity can be made quite narrowband. Since the selectivity is coupled to the LO, a tunable receiver may be achieved that enables selectivity over a wide range of input frequencies.
Full Text FORM 2
THE PATENTS ACT, 1970
(39 of 1970)
&
THE PATENTS RULES, 2003
COMPLETE SPECIFICATION
(Se section 10, rule 13)
MIXER WITH FEEDBACK
TELEFONAKTIEBOLAGET LM ERICSSON (publ), of S-164 83 Stockholm, Sweden
The following specification particularly describes and ascertains the invention and the manner in which it is to be performed.

WO 2005/064786 PCT/EP2004/014517
MIXER WITH FEEDBACK
BACKGROUND OF THE INVENTION
Field of the Invention
5 This invention is directed generally to radio communication systems and
particularly to radio communication systems with integrated receivers. Description of the Related Art
In radio communication systems, a mixer is used to up-convert a baseband signal to a higher frequency (e.g., radio frequency (RF)) signal for ease of transmission. The
10 mixer can also down-convert a high frequency signal to baseband for ease of signal processing. Various types of mixers exist, including unbalanced, single and double balanced, and the four-quadrant or Gilbert mixer. For general information regarding the various types of mixers, the reader is referred to "Radio-Frequency Microelectronic Circuits for Telecommunication Applications," Yannis E. Papananos, ISBN 0-7923-
15 8641-8, Kluwer Academic Publishers, Boston, 1999.
An example of a mixer being employed in a typical receiver 100 is illustrated in Figure 1. The receiver 100 is a homodyne receiver in which an RF signal is converted directly to a baseband signal (in contrast to a heterodyne receiver where the RF signal is first converted to one or more intermediate frequency (IF) signals). As can be seen, the
20 receiver 100 has a number of functional components, including an antenna 102, a low noise amplifier (LNA) 104, a mixer 106, and a local oscillator (LO) 108, the LO 108 typically being a voltage controlled oscillator (VCO). These components are well known to one skilled in the art and will not be described in detail here. Briefly, the antenna 102 receives and provides the RF signal to the receiver 100. The RF signal is then amplified
25 by the low noise amplifier 104 and mixed by the mixer 106 with a signal from the LO 108. The mixing action recovers the baseband signal from the RF signal, which baseband signal is then outputted at the mixer 106. In most instances, the receiver 100 also includes a post mixer amplifier 110 for amplifying the baseband signal and a low pass filter 112 for removing any high frequency component of the baseband signal.
3-
WO 2005/064786 PCT/EP2004/014517
A challenge in modern radio communication systems has been, and continues to be, to design receivers (and transmitters) that can meet increasingly strict performance standards while fitting into ever shrinking packages. To this end, many modern radio receivers (and transmitters) are implemented on a single application specific integrated
5 circuit (ASIC). Integrated radio receivers, however, cannot easily implement RF selectivity due to the very high quality (Q) factors required in modern communication systems. The Q-factor is a ratio of a channel's center frequency over its allowable spread and is a measure of how tightly the frequency of the channel must be controlled. For example, a channel in the Global System for Mobile Communication (GSM) may have a
10 center frequency of 1.8 GHz and may be 200 kHz wide. This corresponds to an extremely high Q-factor of approximately 9000. Band selectivity is also difficult to implement on a chip. For example, a typical cellular band can be some 50Mhz wide and, at a center frequency of 1.8 GHz, corresponds to a relatively high Q-factor of approximately 36.
15 One way to achieve sufficient RF selectivity is to employ surface acoustic wave
(SAW) filters. SAW filters are frequently used in radio communication applications because of their high performance characteristics and low insertion loss. In most receivers, a SAW filter is inserted between the antenna and the low noise amplifier to suppress out-of-band signals. For non-TDMA (time domain multiple access) systems
20 where the receiver and transmitter operate simultaneously, a SAW filter is often inserted between the low noise amplifier and the first down-conversion mixer as well. Adding filters, however, has the drawback of increasing the cost, size, and complexity of the receiver.
Moreover, the problem intended to be solved by the additional filters is to
25 suppress unwanted out-of-band signals sufficiently so that these signals do not desensitize the receiver or cause excessive distortion that may hamper the receiver sensitivity. But desensitization and distortion usually become significant only when the power level at the low noise amplifier input approaches the receiver compression point. The receiver compression point is a figure of merit that indicates how much signal power
30 the receiver can handle before the receiver gain begins to be affected. It is generally

3
WO 2005/064786 PCT/EP2004/014517
considered to be the point where the receiver gain is decreased by 1 dB as a result of an increase in the input power. The reason the gain is affected is because beyond a certain point, the output of the receiver becomes saturated and further increases in input power will not result in corresponding increases in output power. By increasing the receiver
5 compression point significantly, desensitization and distortion may be reduced. As a result, it may be possible to avoid some or all of the additional SAW filters.
Accordingly, what is needed is a way to improve the RF selectivity in an
integrated receiver using few or no additional components, such as the SAW filters
mentioned above. In particular, what is needed is a way to increase the compression
10 point of the receiver, thereby making the receiver less susceptible to desensitization and
distortion effects that may hamper the receiver sensitivity.
SUMMARY OF THE INVENTION
The present invention is directed to method and system for increasing the
15 compression point of a receiver by deriving a feedback signal from mixer output signals. The feedback signal prevents the receiver from going into compression on strong out-of-band or blocking signals, while enhancing the receiver gain at the desired frequency. The desired frequency coincides with the LO signal and is therefore particularly applicable for, but not limited to, homodyne receivers where selectivity can be made quite narrow
20 band. Since the selectivity is coupled to the LO, a tunable receiver may be achieved that enables selectivity over a wide range of input frequencies.
In general, in one aspect, the invention is directed to a method of providing feedback from a mixer to a preceding amplifier in a receiver. The method comprises receiving a radio frequency signal at the radio frequency receiver and generating
25 frequency translated signal from the radio frequency signal. The method further comprises deriving a feedback signal from a combination of mixer output signals, the feedback signal being a function of a frequency of the radio frequency signal. The feedback signal is then provided to the preceding amplifier in the receiver.
In general, in another aspect, the invention is directed to a radio frequency
30 receiver having a mixer feedback. The radio frequency receiver comprises a low noise
WO 2005/064786 PCT/EP2004/014517
amplifier configured to receive a radio frequency signal, the radio frequency signal having a baseband signal carried thereon. The radio frequency receiver further comprises a mixer configured to mix the radio frequency signal with a local oscillator signal to recover the baseband signal. A feedback network connects the mixer to the low noise
5 amplifier and provides a feedback signal to the low noise amplifier, wherein the feedback signal is a function of a frequency of the radio frequency signal.
In general, in yet another aspect, the invention is directed to a radio frequency receiver having feedback from a mixer of the receiver to an input stage of the receiver. The radio frequency receiver comprises an amplifier configured to receive a radio
10 frequency signal, the radio frequency signal having a base band signal carried thereon. The radio frequency receiver further comprises a mixer configured to mix the radio frequency signal with a local oscillator signal, the mixer having at least a high-pass output path and a low-pass output path. A feedback network connects the mixer to the low noise amplifier, the feedback network providing a feedback signal from the high-pass
15 output path to the low noise amplifier.
It should be emphasized that the term comprises/comprising, when used in this specification, is taken to specify the presence of stated features, integers, steps, or components, but does not preclude the presence or addition of one or more other features, integers, steps, components, or groups thereof.
20
BRIEF DESCRIPTION OF THE DRAWINGS
A better understanding of the invention may be had by reference to the following
detailed description when taken in conjunction with the accompanying drawings,
wherein:
25 Figure 1, previously described, is a block diagram of a prior art receiver;
Figure 2 is block diagram of an exemplary receiver according to embodiments of the invention;
Figure 3 is a circuit diagram showing a portion of a mixer; Figure 4 is a circuit diagram showing an exemplary network that may be used to
30 combine the outputs of a mixer according to embodiments of the invention;

WO 2005/064786
1 Figure 5 is a circuit diagram showing a portion of an exemplary receiver having
a feedback network according to embodiments of the invention;
Figure 6 is a circuit diagram showing a portion of an exemplary receiver having a common-base low noise amplifier with shunt feedback according to embodiments of the
5 invention;
Figure 7 is a circuit diagram showing a portion of an exemplary receiver having a common-base low noise amplifier with in-phase and quadrature shunt feedback according to embodiments of the invention;
Figure 8 is a circuit diagram showing a portion of an exemplary receiver having a
10 differential common-base low noise amplifier with mixer shunt feedback according to embodiments of the invention;
Figure 9 is a circuit diagram showing a portion of an exemplary receiver having acommon-emitter low noise amplifier with dual loop mixer feedback according to
embodiments of the invention; and
15 Figure 10 is a circuit diagram showing a portion of an exemplary receiver having
a common-emitter low noise amplifier with a higher-order mixer feedback according to embodiments of the invention.
DETAILED DESCRIPTION OF THE INVENTION
20 Following is a detailed description of the invention with reference to the drawings
wherein reference numerals for the same or similar elements are carried forward. It should be noted that the transistors shown in the drawings are intended to be general in nature and do not indicate a preference for a particular type of transistor. Likewise, the equations provided herein are intended to be general in nature and do not indicate a
25 preference for a specific type of transistor. In addition, all resistors described herein may also be some other form of impedance such as capacitive (C), resistive (R), inductive (L), RC, RL, and the like. In general, the invention is capable of being implemented with any suitable type of transistor (e.g., bi-polar junction transistors (BJT), metal oxide semiconductor field effect transistors (MOSFET), etc.), using any suitable feedback
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mechanism (e.g., capacitive, resistive, inductive, RC, RL, etc.), and using any suitable biasing scheme (e.g., current source, bootstrap, resistors, LC, etc.).
Embodiments of the invention provide a receiver having a significantly improved compression point. The improvement in the receiver compression point is achieved by
5 providing a feedback between the mixer and the low noise amplifier. The mixer, which may be a conventional mixer, has an output that includes a high-pass path and a low-pass path. The low-pass path is fed to the baseband portion of the receiver (i.e., the IF circuitry). The high-pass path, in accordance with embodiments of the invention, is fed back to the low noise amplifier. This feedback reduces the signal swing seen by the
10 devices within the receiver feedback loop and, as a result, increases the compression point for out-of-band signals (by sacrificing circuit gain for these signals).
The feedback causes the low noise amplifier and mixer combination to behave much like an operational amplifier. Specifically, the output signal of the low noise amplifier will be limited by clipping in the mixer, yet the output swing will not be
15 radically changed by the feedback. The mixer output is typically loaded by a filter and therefore will not usually clip, even on strong out-of-band signals, since they will not develop a large voltage swing (provided the bias current is high enough). In contrast, in the non-feedback case, the compression point will be limited by the input (amplifier) stage regardless of the output compression point of the mixer. By adding feedback from
20 the mixer output to the receiver input stage, it is possible to design the feedback network such that its noise contribution is limited (i.e., little local feedback at the input stage) while the compression point will be set by the mixer output current capability (i.e., clipping). Such an arrangement provides more flexibility for circuit designers.
The dual loop feedback (i.e., the combination of mixer feedback and conventional
25 feedback) also enables the control of, for example, the input impedance of the low noise amplifier mixer combination, which would also be useful in reducing the number of matching components. For example, in general, an amplifier block can be described by four transfer parameters: 1/AV, 1/Aj, 1/Gm, and 1/Rm, where Av, Ai, Gra, and Rm are the voltage gain, current gain, transconductance, and transresistance, respectively. By
30 applying one feedback loop to the amplifier, one transfer parameter can be controlled

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(e.g., the voltage gain). By applying two loops, two transfer parameters may be controlled. When both the voltage and current gains are controlled, the input impedance will be defined by their ratio and a fixed, well controlled input impedance is achieved. For simplicity, real values (e.g., conductances) have been used in this example, but in
5 practice, the parameters may assume complex, frequency dependent values. (See, for example, Nordholt, Ernst H., "Design of High-Performance Negative-Feedback Amplifiers," Elsevier, 1983, ISBN 0-444-42140-8; and Davidse, Jan, "Analog Electronic Circuit Design," Prentice Hall, 1991, ISBN 0-13-035346-9.)
Furthermore, in some embodiments, the mixer output stage may be made into a
10 class AB amplifier (i.e., an amplifier wherein the conduction angle is larger than u, but less than 2%), thereby vastly increasing its current drive capability without increasing the average power consumption under normal conditions (much like the output stage in op-amps). (See Davidse, Jan, referenced above for more information regarding class AB amplifiers.) A reasonably high loop gain should be used for optimal performance.
15 Figure 2 illustrates a block diagram of a receiver 200 having a mixer feedback according to the teachings of the invention. As can be seen, the receiver 200 has a feedback 202 from the mixer 106 to the low noise amplifier 104. The purpose of the feedback 202 is to prevent the receiver chain from going into compression on stronger blocking signals while enhancing the gain at the desired frequency. By
20 way of explanation, feedback has traditionally been used in low-frequency applications where no signal frequency translation (i.e., mixing) takes place. In these applications, input and output signals, at least for the blocks enclosed by the feedback loop, have the same frequency. Since the frequencies are the same, the receivers can be more easily designed to have a compression point (and linearity) that exceeds the compression point of a non-
25 feedback system.
For frequency translation systems such as the one shown in Figure 2, however, the input and output signal frequencies of the feedback loop are not the same. As a result, the feedback (i.e., error) signal cannot be derived simply as a scaled down version of the output signal, since the output signal may include components of two or more


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different frequencies. Therefore, in accordance with the teachings of the invention, the feedback signal is instead derived in terms of the frequency of the input signal.
(1) (2) (3)
Figure 3 shows a double-balanced mixer 300 wherein M1-M4 denote transistors that form a mixer core, O1-O4 represent mixer output currents, I1-I2 represent input
5 currents, and S1-S2 represent LO signals. The output currents O1-O4 may be expressed as follows:
0I=(Jo+-icos(a>,O)-S1
^(/o+^cos^OMl-S,)
O3=(/0-^cos(^))-(l-52)
O4=(/0-^cos(©,.0)-52 (4)
10
where Io represents the mixer bias current, Ij and h represent the input signal current plus the mixer bias current (7/+/2=2/o), is represents the mixer signal current, co* represents the input signal frequency, and the input signal currents h and h and LO signal Sj and 5!?,
respectively, can be expressed as O^2=Ih2-Sh2=I0±—cos(a>it)-Sl2, JSJ = 1 — 15 After appropriate substitution and simplification, Equations (l)-(4) maybe rewritten as:

0,«

J0+-icos(a>,0

- + -cos(c»00 + ...
Z 7t



T O T
— + -£-cos(0,f) + —-cos(ay) +
4* *T 7b

(5)



-?- [cos(A0 + Aw)t)] 2TC

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I0 +^-cos(a>,0


12 , A
cos(«00-
2 7T
7" * O 7"
— +—cos(fij;0 —— cos(o00 -
7T

(6)

-^-jcosCAatf) + cos((2tf>0 + Aw)t)] In

O,*

ro —-cos(a>,*)

-—cos(fl>00-... =



T or
— —-cos(#,/) °-cos(o)Qt) +
2 4 n

(J)

-^-[cos(A^) + cos((2fi?0 +Aw)0]
2#

70-^-cos(©,0


- + -cos(fi>00 + -.. 2 #
T * or
-5- - ^-COS(fi>,0 + —S-COS^oO -
2 4 TT
—[cos(Aftrf) + cos((2«0 + Aw)t)] In

(8)

In most receivers, the baseband output 0BB is derived as a linear combination of the output currents in Equations (5)-(8), as follows:

2z
0BB = O, + 03 - (02 + 04) = —*-[cos(Aort) + cos((2 7t

(9)

10 where the Aco term corresponds to a down converted signal and the 2COQ term corresponds
to an up converted signal, which can easily be removed with a filter, since Aoo and 2CDO
are typically widely separated in frequency.
Note in Equation (9) that there is a total absence of any input signal frequency
term. As a result, the traditional baseband mixer output OBB is not feasible as a feedback
15 signal to the low noise amplifier. By combining the mixer output currents expressed in
Equations (5)-(8) in a certain way, however, the mixer output can be derived in terms of
the input signal frequency, as follows:

WO 2005/064786 PCTYEP2004/014517
* ORF=O1+O2-(O3+O4) = IS cos (wit (10)
where ORF is the RF output of the mixer. Note the presence of the input signal frequency term ecu in the Equation (10). Thus, by combining the mixer output currents in accordance with teachings of the invention, the mixer output may be derived as a
5 function of the input signal frequency ©,-. And since Aou and cot are typically widely separated in frequency, both OBB and ORF may be generated.
One way to combine the mixer output currents to achieve the above result is by using, for example, a simple passive network. Figure 4 illustrates an exemplary passive network 400 that can be used to generate the RF output, ORF, of the mixer. The network
10 400 includes resistors R1-R4 and capacitors C1-C4, all interconnected as shown. Applying well known circuit analysis techniques, it can be shown that Equation (10) may be implemented by tapping a connection between CI & C2 and C3 & C4 to obtain the mixer RF output ORF.
An example of how the passive network 400 may be implemented in a mixer is
15 shown in Figure 5, where bias and other details have been omitted for ease of illustration. Figure 5 illustrates a typical single-balanced mixer 500 that has been modified in accordance with the teachings of the present invention. Note that although a single-balanced mixer is shown, one skilled in the art may readily expand the concepts herein to include double-balanced mixers, four-quadrant mixers, and other types of mixers.
20 The single-balanced mixer 500 includes transistors Ql & Q2 that together form
the mixer core, and transistors Qx & Qy that together form the low noise amplifier. Resistors Rl & R2 are connected between the collectors of Ql & Q2 and the voltage supply Vcc. The resistors Rl & R2 are output resistors and correspond to resistors Rl & R2 in Figure 4. Capacitors CI & C2 are connected between the collectors of Ql & Q2
25 and the emitter of Qx. The capacitors CI & C2 correspond to capacitors CI & C2 in Figure 4 and together form a mixer feedback 502. VBB is a baseband output of the mixer, VLO is an LO signal, and VRF is an RF input to a low noise amplifier.
In operation, the mixer signal current is will asymptotically approach:
(II)

WO 2005/064786 PCT/EP2004/014517
when the loop gain increases. As 4 does so, the compression point will begin to be
controlled by clipping in the mixer signal current is and not by any input device non-linearity. Thus, it becomes possible to decouple the compression point from the operation of the input device. Since the feedback capacitors CI & C2 will approximate
5 short circuits at high frequencies (e.g., RF), the baseband output will have a common-mode component equal to VRF, which is tractable for most blocking requirements. In some embodiments, additional low-pass filtering of VBB may be implemented for improved performance.
Since Equation (11) is valid for reasonably high loop gains (e.g., above 10-20dB),
10 taking Equation (9) into consideration, the baseband output VBB can be written as:
(12)

where RBB is the low frequency loading on the mixer output (which can be of much
2 higher impedance than the high frequency loading). The factor — is a result of the
frequency translation process, but otherwise the baseband output VBB is proportional to
15 the input signal (minus the desired frequency shift).
An advantage of the above mixer arrangement is the bandwidth of VBB is
primarily limited by the tuning range of the LO (e.g., a VCO) and the low noise amplifier
2 R
input match. The selectivity is approximately equal to the baseband gain, or —, for
it Re
the configuration of Figure 5 before any other filtering is considered. Similar expressions
20 may be developed for other configurations. Another advantage is that the mixer may be
implemented entirely as an integrated circuit (i.e., no non-ASIC components are
required).
Following are exemplary implementations of other types of mixers that may be
used in accordance with embodiments of the invention. Persons having ordinary skill in
25 the art will recognize the advantages and benefits these various exemplary
implementations. Figure 6, for example, illustrates a common-base low noise amplifier


WO 2005/064786
with shunt feedback mixer 600 that is similar to the mixer of Figure 5, except that the
feedback capacitors CI & C2 are connected to the collector of transistor Qx.
Figure 7 illustrates an exemplary mixer 700 according to embodiments of the invention, implemented using a common-base low noise amplifier with in-phase and
5 quadrature shunt feedback. The mixer 700 in Figure 7 is similar to the mixer 600 of Figure 6, except the local oscillator signal Vw is applied to both the in-phase (Ql & Q2) and quadrature (Q3 & Q4) inputs of the mixer. In Figure 7, feedback resistors R3 & R4 as well as capacitors C3 & C4 perform similar functions as their counterparts Rl & R2 andCl&C2.
10 Figure 8 illustrates an exemplary mixer 800 according to embodiments of the
invention, implemented using a differential common-base low noise amplifier with mixer shunt feedback. As can be seen, the mixer 800 of Figure 8 is a balanced version of the mixer 600 of Figure 6 (where subscripts "a" and "b" denote the two balanced paths), with a balanced low noise amplifier and a double-balanced mixer. Here, the term double-
15 balanced refers to both the RF input and the baseband output being balanced, as opposed to the mixer 600 in Figure 6, which has a single-balanced mixer with one RF input and a balanced baseband output.
Figure 9 illustrates an exemplary mixer 900 according to embodiments of the invention, implemented using a common-emitter low noise amplifier with dual loop
20 mixer feedback. As can be seen, the mixer in Figure 9 includes a low noise amplifier composed of Qx, Qz, and Qy together with a single-balanced mixer core Ql & Q2. The mixer 900 has two feedback loops, with CI & C2 and Re setting the voltage gain, and Rf and R3 setting the current gain. As a result, the input impedance of the mixer 900 will be defined by the two loops when the loop gains are high. Assuming high loop gains, the
25 input impedance can be approximated as Zin=Re(l+Rf/R3) for RF frequencies.
Figure 10 illustrates an exemplary mixer 1000 according to embodiments of the invention, implemented using a common-emitter low noise amplifier with a higher-order mixer feedback. In Figure 10, resistor R3 and capacitor C3 add another high-pass pole to the feedback network, making the cut-off slope steeper. That is, the attenuation of the
30 signal changes more rapidly with frequency. Similarly, resistors R4 & R5 and capacitors

WO 2005/064786 PCT/EP2004/014517
C4 & C5 add another low-pass pole at the baseband output to increase the suppression
of the RF signal at the mixer output. Thus, by using a higher-order mixer feedback,
better selectivity may be achieved. But there may be some limitation due to stability
constraints and, therefore, care has to be exercised when using this implementation.
5 In addition to the foregoing embodiments, other combinations of feedback
structures, including multi-loop feedback structures with a mix of pre-mixer and post-mixer feedback, are also possible. Furthermore, both first order networks and higher order networks may also be used. And while embodiments of the invention have been described with respect to an integrated receiver, the teachings of the present invention
10 may be readily applied to non-integrated receivers as well.
Thus, while particular embodiments and applications of the present invention have been illustrated and described, it is to be understood that the invention is not limited to the precise construction and compositions disclosed herein, and that modifications and variations may be made to the foregoing without departing from the scope of the
15 invention as defined in the appended claims.
43

WO 2005/064786 PCT/EP2004/014517

We Calaim

1. A method of providing feedback from a mixer to a preceding amplifier in a receiver, comprising:
receiving a radio frequency signal at the radio frequency receiver;
5 generating a frequency translated signal from the radio frequency signal;
deriving a feedback signal from a combination of mixer output signals, the feedback signal being a function of a frequency of the radio frequency signal; and providing the feedback signal to the preceding amplifier in the receiver.
10 2. The method according to claim 1, wherein the feedback signal is derived
from a feedback network connected to the mixer, the feedback network including frequency selective elements capable of separating out the feedback signal from an output of the mixer.
15 3. The method according to claim 2, wherein the feedback network is a first
order network.
4. The method according to claim 2, wherein the feedback network is a
higher order network.
20
5. The method according to claim 2, wherein the feedback network uses
single-loop feedback.
6. The method according to claim 2, wherein the feedback network uses
25 multiple feedback loops, and at least one feedback loop provides the feedback signal to the preceding amplifier
7. The method according to claim 1, wherein the mixer is a single-balanced
mixer.
30


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8. The method according to claim 1, wherein the mixer is a double-balanced mixer.
9. The method according to claim 1, wherein the mixer is a quadrature mixer.
5
10. The method according to claim 1, wherein the preceding amplifier is a low
noise amplifier and the feedback signal is provided to the low noise amplifier.
11. A radio frequency receiver having a mixer feedback, the radio frequency
10 receiver comprising:
a low noise amplifier configured to receive a radio frequency signal, the radio frequency signal having a baseband signal carried thereon;
a mixer configured to mix the radio frequency signal with a local oscillator signal
to recover the baseband signal; and
15 a feedback network connecting the mixer to the low noise amplifier, the feedback
network providing a feedback signal to the low noise amplifier, wherein the feedback signal is a function of a frequency of the radio frequency signal.
12. The receiver according to claim 11, wherein the feedback network
20 includes frequency selective elements capable of separating out the feedback signal from
an output of the mixer.
13. The receiver according to claim 11, wherein the feedback network is a
first order network.
25
14. The receiver according to claim 11, wherein the feedback network is a
higher order network.
15. The receiver according to claim 11, wherein the feedback network
30 comprises a single-loop feedback.
AS* I*

WO 2005/064786
16. The receiver according to claim 11, wherein the feedback network
comprises a multiple feedback loops, and at least one feedback loop provides the
feedback signal to the low noise amplifier.
5
17. The receiver according to claim 11, wherein the mixer is a single-balanced
mixer.
18. The receiver according to claim 11, wherein the mixer is a double-
10 balanced mixer.
19. The receiver according to claim 11, wherein the mixer is a quadrature
mixer.
15 20. The receiver according to claim 11, wherein receiver is integrated on a
single application-specific integrated circuit (ASIC).
21. A radio frequency receiver having feedback from a mixer of the receiver
to an input stage of the receiver, the radio frequency receiver comprising:
20 an amplifier configured to receive a radio frequency signal, the radio frequency
signal having a baseband signal carried thereon;
a mixer configured to mix the radio frequency signal with a local oscillator signal, the mixer having at least a high-pass output path and a low-pass output path; and
a feedback network connecting the mixer to the low noise amplifier, the feedback 25 network providing a feedback signal from the high-pass output path to the low noise amplifier.
22. The radio frequency receiver according to claim 21, wherein the receiver
is implemented using a common-base low noise amplifier with shunt feedback and the
30 feedback network is connected to an emitter of the low noise amplifier.
46 II-

WO 2005/064786 PCT/EP2004/014517

10


23. The radio frequency receiver according to claim 21, wherein the receiver is implemented using a common-base low noise amplifier with shunt feedback and the feedback network is connected to a collector of the low noise amplifier.
24. The radio frequency receiver according to claim 21, wherein the receiver is implemented using a common-base low noise amplifier with in-phase and quadrature shunt feedback and the local oscillator signal is applied to in-phase and quadrature inputs of the mixer.
25. The radio frequency receiver according to claim 21, wherein the receiver is implemented using a differential common-base low noise amplifier with mixer shunt feedback.

15 26. The radio frequency receiver according to claim 21, wherein the receiver
is implemented using a common-emitter low noise amplifier with dual loop mixer feedback.
27. The radio frequency receiver according to claim 21, wherein the receiver 20 is implemented using a common-emitter low noise amplifier with a higher-order mixer feedback.
28. A method of providing feedback from a mixer to a preceding amplifier in a receiver, substantially as herein described with reference to the accompanying drawings.

29.

A radio frequency receiver having a mixer feedback, substantially as herein described with reference to the accompanying drawings.
Dated this 3rd Day of July, 2006
G. Deepak Sriniws ———-> OfK&S Partners Agent for the Applicants.

Abstract:
A method and system for increasing the compression point of a receiver by deriving a feedback signal from mixer output signals. The feedback signal prevents the receiver from going into compression on strong out-of-band or blocking signals, while enhancing the receiver gain at the desired frequency. The desired frequency coincides with the local oscillator (LO) signal and is therefore particularly applicable for, but not limited to, homodyne receivers where selectivity can be made quite narrowband. Since the selectivity is coupled to the LO, a tunable receiver may be achieved that enables selectivity over a wide range of input frequencies.
-IQ.

WO 2005/064786

PCT/EP2004/014517

1/6
J 00

>K
102

FIG. 1
(PRIOR ART)

>K
102

ft DKEFAK SRtNfWAf
OF K & & Partner*

n

WO 2005/064786

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2/6
30D


0,1 02 03 ]0A

FIG. 3

4
/
> R4
HH
003
R1> r r > *2 Rs| r r
.Ct C2> > C3 C4
lh
)r"HH

400



OBB

;

0RF

1

J7G. 4
OF K & & Partner* i
AGENT FOR THE AW»UCXifl»

WO 2005/064786

PCT/EP2004/014517

3/6

500 v

"Vcc



*c

Ri:

VflB

VRF —(v

Ci
Qx

02

|-(C01 wty-\_

^iO

c=t= >R
J, FIG, 5


Ks^
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The undersigned certifies that attached hereto is a true and accurate copy of the Assignment of the invention for MIXER WITH
FEEDBACK, by Sven Mattisson dated May 10, 2004.
IN WITNESS WHEREOF, I have hereunto set my hand and seal this 20th day of June 2006.



Sharon D. Gamble Paralegal

STATE OF TEXAS COUNTY OF COLLIN

§ ss,



On this 20th June 2006, before me, a Notary Public in and for the State and County aforesaid, personally appeared Sharon D. Gamble, to me well known and known by me to be the person of the above name who signed and sealed the foregoing instrument, and aalf.nrmil arifrariL tha.swtaa. fap be her own free act and deed.
J^KS&L PAMELA KAY EWING
nUUl Notary Public, State of Texas
'"*>/•./ My Commission Expires
.Rwr November 14,2006
[
■ia-Mni ■ ■ ■ ■ '■ ■ -
Notary Public in and for the State of Texas


Pamela K. Ewing

EUS/J/P-03:8192

Documents:

781-MUMNP-2006-ABSTRACT(11-4-2011).pdf

781-mumnp-2006-abstract(5-7-2006).pdf

781-mumnp-2006-abstract-1.jpg

781-mumnp-2006-abstract.doc

781-mumnp-2006-abstract.pdf

781-mumnp-2006-calims.doc

781-mumnp-2006-claims(5-7-2006).pdf

781-MUMNP-2006-CLAIMS(AMENDED)-(11-4-2011).pdf

781-MUMNP-2006-CLAIMS(AMENDED)-(22-9-2011).pdf

781-MUMNP-2006-CLAIMS(MARKED COPY)-(11-4-2011).pdf

781-MUMNP-2006-CLAIMS(MARKED COPY)-(22-9-2011).pdf

781-mumnp-2006-claims.pdf

781-MUMNP-2006-CORRESPONDENCE(22-9-2011).pdf

781-mumnp-2006-correspondence(25-10-2007).pdf

781-mumnp-2006-correspondence-others.pdf

781-mumnp-2006-description (complete).pdf

781-mumnp-2006-description(complete)-(5-7-2006).pdf

781-MUMNP-2006-DRAWING(11-4-2011).pdf

781-mumnp-2006-drawing(5-7-2006).pdf

781-MUMNP-2006-FORM 1(11-4-2011).pdf

781-mumnp-2006-form 18(25-10-2007).pdf

781-mumnp-2006-form 2(5-7-2006).pdf

781-MUMNP-2006-FORM 2(TITLE PAGE)-(11-4-2011).pdf

781-mumnp-2006-form 2(title page)-(5-7-2006).pdf

781-MUMNP-2006-FORM 26(11-4-2011).pdf

781-MUMNP-2006-FORM 3(11-4-2011).pdf

781-mumnp-2006-form 3(18-8-2006).pdf

781-MUMNP-2006-FORM 3(22-9-2011).pdf

781-MUMNP-2006-FORM PCT-IB-304(11-4-2011).pdf

781-mumnp-2006-form-1.pdf

781-mumnp-2006-form-2.doc

781-mumnp-2006-form-2.pdf

781-mumnp-2006-form-3.pdf

781-mumnp-2006-form-5.pdf

781-MUMNP-2006-PETITION UNDER RULE 137(11-4-2011).pdf

781-MUMNP-2006-PROCECUTION HISTORY OF CORRESPONDING APPLICATION(11-4-2011).pdf

781-MUMNP-2006-REPLY TO EXAMINATION REPORT(11-4-2011).pdf

781-MUMNP-2006-REPLY TO EXAMINATION REPORT(22-9-2011).pdf

781-MUMNP-2006-REPLY TO HEARING(7-10-2011).pdf

781-mumnp-2006-wo international publication report(5-7-2006).pdf

abstract1.jpg


Patent Number 249169
Indian Patent Application Number 781/MUMNP/2006
PG Journal Number 42/2011
Publication Date 21-Oct-2011
Grant Date 08-Oct-2011
Date of Filing 05-Jul-2006
Name of Patentee TELEFONAKTIEBOLAGET LM ERICSSON (PUBL)
Applicant Address S-164 83 STOCKHOLM SWEDEN
Inventors:
# Inventor's Name Inventor's Address
1 MATTISSON, SVEN OSTANVAG 3, S-237 36 BJARRED, SWEDEN
PCT International Classification Number H03D7/14
PCT International Application Number PCT/EP2004/014517
PCT International Filing date 2004-12-21
PCT Conventions:
# PCT Application Number Date of Convention Priority Country
1 10/746,330 2003-12-23 U.S.A.