| Title of Invention | A CODE DIVISION MULTIPLE ACCESS (CDMA) TRANSMISSION SYSTEM |
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| Abstract | ABSTRACT A system for communicating information signals using spread spectrum communication techniques. . PN sequences are constructed that provide orthogonality between the users so that mutual interference will be reduced, allowing higher capacity and better link performance. With orthogonal PN codes, the cross-correlation is zero over a predetermined time interval, resulting in no interference between the orthogonal codes, provided only that the code time frames are time aligned with each other. In an exemplary embodiment, signals are communicated between a cell-site and mobile units using dirept sequence spread spectrum communication signals. In the cell-to-mobile link, pilot, sync, i paging and voice channels are defined. Information communicated \. on the cell-to-mobile link channels are, in general, encloded, interleaved, bi phase shift key (BPfK) modulated with orthogonal . covering of each BPSK symbol along with quadrature phase shift key (QPSK) spreading of the covered symbols. In the mobile-to- [ - . cell link, access and voice channells are defined. Information communicated on the mobile-to-cell link channels are, in general, encoded, interleaved, orthogonal | signalling along with QPSK . spreading. ' |
| Full Text | I The present invention relates to cellular telephone systems. More specifically, the present invention relates to a novel and improved system and method for communicating information, in a mobile cellular telephone system or satellite mobile telephone system, using spread spectrum communication signals. The use of code division multiple access (CDMA) modulation techniques is one of several techniques for facilitating communications in which a large number of system users are present. Other multiple ac In the just mentioned patent, a multiple access technique is disclosed where a large number of mobile telephone system users each having a transceiver communicate through satellite repeaters or terrestrial base stations (also referred to as cell-sites stations, cell-sites or for short, cells) using code division multiple access (CDMA) spread spectrum communication signals. In using CDMA communications, the frequency spectrum can be reused multiple times thus permitting an increase in system user capacity. The use of CDMA results in a much higher spectral efficiency than can be achieved using other multiple access techniques. The satellite channel typically experiences fading that is characterized as Ridan. Accordingly the received signal consists of a direct component summed with a multiple reflected component having Rayleigh fading statistics. The power ratio between the direct and reflected component is typically on the order of 6-10 dB, depending upon the characteristics of the mobile unit antenna and the environment about the mobile unit. Contrasting with the satellite channel, the terrestrial channel experiences signal fading that typically consists of the Rayleigh faded component without a direct component. Thus, the terrestrial channel presents a more severe fading environment than the satellite channel in which Rician fading is the dominant fading characteristic. The Rayleigh fading characteristic in the terrestrial channel signal is caused by the signal being reflected from many different features of the physical environment. As a result, a signal arrives at a mobile unit receiver from many directions with different transmission delays. At the UHF frequency bands usually employed for mobile radio communications, including those of cellular mobile telephone systems, significant phase differences in signals traveling on different paths may occur. Tlie possibility for destructive summation of the signals may result, with on occasion deep fades occurring. Terrestrial channel fading is a very strong function of the physical position of the mobile unit. A small change in position of the mobile unit changes the physical delays of all the signal propagation paths, which further results in a different phase for each path. Thus, the motion of the mobile unit through the environment can result in quite rapid fading process. For example, in the 850 MHz cellular radi frequency band, this fading can typically be as fast as one fade per secon per mile per hour of vehicle speed. Fading this severe can be extremel disruptive to signals in the terrestrial channel resulting in poc communication quality. Additional transmitter power can be used t overcome the problem of fading. However, such power increases effe The CDMA modulation techniques disclosed in U.S. Patent Nc 4,901,307 offer many advantages over narrow band modulatioi techniques used in communication systems employing satellite o terrestrial repeaters. The terrestrial channel poses special problems t( any communication system particularly with respect to multipatl signals. The use of CDMA techniques permit the special problems of th( terrestrial channel to be overcome by mitigating the adverse effect o multipath, e.g. fading, while also exploiting the advantages thereof. In a CDMA cellular telephone system, the sam.e frequency banc can be used for communication in all cells. The CDMA wavefornr properties that provide processing gain are also used to discrirninatt between signals that occupy the same frequency band. Furthermore the high speed pseudonoise (PN) modulation allows many different propagation paths to be separated, provided the difference in path delays exceed the PN chip duration, i.e. 1/bandwidth. If a PN chip rate ol approximately 1 MHz is employed in a CDMA system, the full spread spectrum processing gain, equal to the ratio of the spread bandwidth tc system data rate, can be employed against paths that differ by more than one microsecond in path delay {torn the desired path. A one microsecond path delay differential! corresponds to differential path distance of approximately 1,000 feet.; The urban environment typically provides differential path delays in excess of one microsecond, and up to 10-20 microseconds are reported in some areas. In narrow band modulation systems such as the analog FM modulation employed by conventional telephone systems, the existence of multiple paths results in severe multipath fading. With wide band CDMA modulation, however, the different paths may be discriminated against in the demodulation process. This discrimination greatly reduces the severity of multipath fading. Multipath fading is not totally eliminated in using CDMA discrimination techniques because there will occasionally exist paths with delayed differentials of less than the PN chip duration for the particular system. Signals having path delays on this order cannot be discriminated against in the demodulator, resulting in some degree of fading. It is therefore desirable that some form of diversity be provided which would permit a system to reduce fading. Diversity is one approach for mitigating the deleterious effects of fading. Three major types of diversity exist: time diversity, frequency diversity and space diversity. Time diversity can best be obtained by the use of repetition, lime interleaving, and error detection and coding which is :; form of repetition. The present invention employes each of these techniques as a form of time diversity. . CDMA by its inherent nature of being a wideband signal offers a form of frequency diversity by spreading the signal energy over a wide bandwidth. Therefore, frequency selective fading affects only a small part of the CDMA signal bandwidth. Space or path diversity is obtained by providing multiple signal paths through simultaneous links from a mobile user through two or more cell-sites. Furthermore, path diversity may be obtained by exploiting the multipath environment through spread spectrum processing by allowing a signal arriving with different propagation delays to be received and processed separately. Examples of path diversity are illustrated in copending U.S. Patent Application entitled "SOFT HANDOFF IN A CDMA CELLULAR TELEPHONE SYSTEM", Serial No. 07/433.030, filed November 7, 1989, and copending U.S. Patent Application entitled "DIVERSITY RECEIVER IN A CDMA CELLULAR TELEPHONE SYSTEM", Serial No. 07/432,552, also filed November 7, 1989, both assigned to the assignee of the present invention. The deleterious effects of fading can be further controlled to a certain extent in a CDMA system by controlling transmitter power. A system for cell-sile and mobile unit power control is tlisclosed in copending U.S. Patent Application entitled "MI-THOD AND APPARATUS I-OK CONTROLLING TRANSMISSION POWER IN A CDMA CELLULAR MOBILE TELEPHONE SYSTEM", Serial No. 07/433,031, filed November 7, 1989, also assigned to the assignee of the present invention. The CDMA techniques as disclosed in U.S. [\itenl No. 4,901,307 contemplated the use of coherent modulation and demodulation for both directions of the link in mobile-satellite communications. Accordingly, disclosed therein is the use of a pilot carrier signal as a (oiu'rent phase reference for the salellile-to-mobil The CDMA techniques as disclosed in U.S. Patent No. 4,901,307 furtlier contemplated the use of relatively long PN sequences with each user channel being assigned a different PN sequence. The cross-correlation between different PN sequences and the autocorrelation of a r^N sequence for al! time shifts other than zero both have a zero average value which allows the different u.ser signals to l»e discriminated upon reception However, such PN signals are not orthogonal. Although the cross-correlations average to zero, for a short lime interval such as an information bit time the cross-correlation follows a binomial distribution. As such, the sit»nnls inli'rf«>rp wilh oach olhor mnrh tho same as if they were wide bandwidth Gaussian noise at the same power spectral density. Thus the other user signals, or mutual interference noise, ultimately limits the achievable capacity. The existence of multipath can provide path diversity to a wideband PN CDMA system. If two or more paths are available with greater than one microsecond differential path delay, two or more PN receivers can be employed to separately receive these signals. Since these signals will typically exhibit independence in multipath fading, i.e., they usually do not fade together, the outputs of the two receivers can be diversity combined. Therefore a loss in performance only occurs when IxUh receivers experience fades at the same time. Hence, one aspect of (lie present invention is the provisuMi of two or more PN receivers in ciMiibination with a diversity combiner. In order to exploit the existence of multipath signals, to overcome fading, it is necessary to utilize a waveform that permits path diversity combining operatitins to be performed. It is therefi>re an object of the! prest'iit invention to prt)vide for tiie generation of PN sequences which are orthogonal so as to reduce mutual interference, thereby permitting greater user capacity, and sup{>ort [i.ith diversity fhereby overcoming fading. SUMMARY OF THE INVENTION The implementation of spread spectrum communication techniques, particularly CDMA technii]ues, in the mobile celiuKir telephone environment therefore: provides fe.ilures which v.islly enhance system reliability and capacity over other communication system techniques CDMA techniques .is previously mentioned further enable problems such as fading and interference to be readily overcome Accordingly, CDMA techniques further promote greater frequency reuse, thus enabling a substantial increase in the number of system users. The present invention is a novel and improved method and system for constructing PN sequences that provide orthogonality between the users so that mutual interference will be reduced, allowing higher capacity and better link performance. With orthogonal PN codes, the cross-correlation is zero over a predetermined time interval, resulting in no interference between the orthogonal codes, provided only that the code time frames are time aligned with each other. In an exemplary embodiment, signals are communicated between a cell-site and mobile units using direct sequence spread spectrum communication signals. In the cell-to-mobile link, pilot, sync, paging and voice channels are defined. Information communicated on the cell-to-mobile link channels are, in general, encoded, interleaved, bi-phase shift key (Bl^K) modulated with orthogonal covering of each BPSK symbol along with quadrature phase shift key (QPSK) spreading of the covered symbols. In the mobile-to-cell link, access and voice channels are defined. Information communicated on the mobile-to-ccU link channels are, in general, encoded, interleaved, orthogonal signalling along with QPSK spreading. BRIEF DESCRIPTION OF THE DRAWINGS The features, objects, and advantages of the present invention will become more apparent from the detailed description set forth below when taken in conjunction with the drawings in which like reference characters identify correspondingly thrt)ughout and wherein: Figure 1 is a schematic overview of an exemplary CDMA cellular lolephone system; Figure 2 is a block diagramj of the cell-site equipment as implemented in the CDMA cellular telephone system; Figure 3 is a block diagram of the cell-site receiver; Figure 4 is a block diagram of the cell-site transmit modulator; and * Figure 5 is an exemplary timing diagram of sync channel symtx^I synchronization; Figure 6 is an exemplary liming diagram of sync channel timing with orthogonal covering; Figure 7 is an exemplary liming diagram of the overall cell-to-mobile link timing; Figure 8 is a block diagram of the mobile telephone switching office equipment; Figure 9 is a block diagram of the mobile unit telephone configured for CDMA communications in the CDMA cellular telephone system; Figure 10 is a block diagram of the mobile unit receiver; and Figure 11 is a block diagram of the mobile unit transmit modulator; Figure 12 is an exemplary liming diagram of the mobile-to-ccU link for the variable data rale with burst transmission; and I"iguro 13 is an exemplary timing ciiagram of the t>verall mobile-to-celi link timing. DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS In a CDMA cellular telephone system, each cell-site has a plurality of modulator-demodulator units or spread spectrum modems. Each mcxiem consists of a digital spread spectrum transmit modulator, at least one digital spread spectrum data receiver and a searcher receiver. Fach HKxiem at the cell-site is assigned to a mobile unit as needed to facilitate communications with the assigned mobile unit. A soft handoff scheme is employed for a CDMA cellular telephone system in which a new cell-site modem is assigned to a mobile unit while the old cell-site modem continues to service the call. When the mobile unit is located in the transition region between the two cell-sites. the call can be switched back and forth between cell-sites as signal strength dictates. Since the mobile unit is always communicating through at least one cell-site modem, fewer disrupting effects to the mobile unit or in service will occur. The mobile unit thus utilizes multiple receivers for assisting in the handoff process in addition to a diversity function for mitigating the effects of fading. In the CDMA cellular telephone system, each cell-site transmits a "pilot carrier" signal. Should the cell be divided into sectors, each sector has an associated distinct pilot signal within the cell. TTiis pilot signal is used by the mobile units to obtain initial system synchronization and to provide robust time, frequency and phase tracking of the cell-site transmitted signals. Each cell-site also transmits spread spectrum modulated information, such as cell-site identification, system timing, mobile paging information and various other control signals. The pilot signal transmitted by each sector of each cell is of the same spreading code but with a different code phase offset. Phase offset allows the pilot signals to be distinguished from one another thus distinguishing originating cell-sites or sectors. Use of the same pilot signal code allows the mobile unit to find system timing synchronization by a single search through all pilot signal code phases. The strongest pilot signal, as determined by a correlation process for each code phase, is readily identifiable. The identified strongest pilot signal generally corresponds to the pilot signal transmitted .by the nearest Cell-site. However, the strongest pilot signal is used whether or not it is transmitted by the closest cell-site. Upon acquisition of the strongest pilot signal, i.e. initial synchronization of the mobile unit with the strongest pilot signal, the mobile Unit searches for another carrier intended to bo received by all system users in the cell. Tliis carrier, cialled the synchronization channel, transmits a broadcast message containing system information for use by the mobiles in the system. The system information identifies the cell-site and the system in addition to conveying information which allows the long PN codes, interleaver frames. Vocoders and other system timing information used by the mobile mobile unit to be synchronized without additional searching. Another channel, called the paging channel may also be provided to transmit messages to mobiles indicating that a call , has arrived for them, and to respond with channel assignments when a mobile initiates a call. The mobile unit continues to scan the received pilot carrier signal code at the code offsets corresponding to cell-site neighboring sector or neighboring transmitted pilot signals. This scanning is done in order to determine if a pilot signal emanating from a neighboring sector or cell is becoming stronger than the pilot signal first determined to be strongest. If, while in this call inactive mode, a neighbor sector or neighbor cell-site pilot signal becomes stronger than that of the initial cell-site sector or cell-site transmitted pilot signal, the mobile unit will acquire the stronger pilot signals and corresponding sync and paging channel of the now sector or cell-site. When a call is initiated, a psoudonoise (I'N) code address is determined for use during the course of this call. The code address may be either assigned by the cell-site or be determined by prearrangement based upon the identity of the moDiie unit. After a call is initiated the mobile unit continues to scan the pilot signal transmitted by the cell-site through which communications are established in addition to pilot signal of neighboring sectors or cells. Pilot signal scanning continues in order to determine if one of the neighboring sector or cell transmitted pilot signals becomes stronger than the pilot signal transmitted by the cell-site the mobile unit is in communication with. When the pilot signal associated with a neighboring cell or cell sector becomes stronger than the pilot signal of the current cell or cell .sector, it is an iiuiicalion to the mobile unit that a new cell or cell sector has b«^en entered and that a handoff should be initiated. An exemplary telephone system in which the present invention is embodied is illustrated in Figure 1. The system illustrated in Figure 1 utilizes spread spectrum modulation techniques in communication between the system mobile units or rrtobile telephones, and the cell-sites. Cellular systems in large cities may have hundreds of cell-site stations serving hundreds of thousands of mobile telephones. The use of spread sp«>ctrum techniques, in particular CDMA, readily facilitates iiuroasos in user capacity in systems of this size as compared to conventional FM iTioduialion cellular systems. In Figure 1, system controller and switch TO, also referred to as mobile telephone sw^itching office (MTSO), typically includes interface and processing circuitry for providing system control to the cell-sites. Controller 10 also controls the routing of telephone calls from the public switched telephone network (PSTN) to the appropriate cell-site for transmission to the appropriate mobile unit. Controller 10 also controls the routing of calls from the mobile units, via at least one cell-site, to the PSTN. Controller 10 may connect calls between mobile users via the appropriate cell-sites since the mobile units do not typically communicate directly with one another. Controller 10 may be coupled to the cell-sites by various means such as dedicated telephone lines, optical fiber links or microwave communication links. In Figure 1, two such exemplary ceU-sites 12 and 14 includint>, along with mobile units 16 and 18 each including a cellular telephone are illustrated. Cell-sites 12 and 14 as discussed herein and illustrated in the drawings are considered to service an entire cell. I lowever it should be understood that the cell may be geographically divided into sectors with each sector treated as a different coverage area. Accordingly, handoffs are made between sectors of a same cell as is described herein for multiple cells, while diversity may also be achieved iKHween sectors as is for cells. In Figure 1, arrowed lines 20a-20b and 22a-22b respectively define the possible communication links between cell-site 12 and mobile unit 16 and 18. Similarly, arrowed lines i24a-24b and 26a'26b respectively define the possible communication links between cell-site 14 and mobile units 16 and 18. Cell-sites 12 and 14 nominally transmit using equal power. The cell-site service areas or cells are designed in geographic shapes such that the mobile unit will normally be closest to one cell-site, Should a call be initiated within the PSTN, controller 10 transmits the call information to all the cell-sites in tlie area. The cell-sites in ret\irn transmit a pagi-ng message within each respective coverage area that is intended for the called recipient mobile user. When the intended recipient mobile unit hears the page message, it responds with a control message ..that is transmitted to the nearest cell-site. This control message signals the system controller that this particular cell-site is in communication with the mobile unit; Controller 10 then routes the call through this cell-site to the mobile unit. Should mobile unit 16 move out of the coverage area of the initial cell-site, cell-site 12, an attempt is made to continue the call by routing the call through another cell-site. With respect to cellular t other chosen by lottery. Because of the order in which allocations were made, the 12.5 MHz allocated to each carrier for each direction of the link is further subdivided into two sub-bands. For the wireline carriers, the sub-bands are each 10 MHz and 2.5 MHz wide. For the non-wireline carriers, the sub-bands are each 11 MHz and 1.5 MHz wide. Tlius, a signal bandwidth of less than 1.5 MHz could be fit into any of the sub-bands, while a bandwidth of less than 2.5 MHz could be fit into all but one sub-band. To preserve maximum flexibility in allocating the CDMA technique to the available cellular frequency spectrum, the waveform utilized in the cellular telephone system should be less than 1.5 MHz in bandwidth. A good second choice would be a bandwidth of about 2.5 MHz, allowing full flexibility to the wireline cellular carriers and nearly full flexibility to non-wireline cellular carriers. While using a wider bandwidth has the advantage of offering increased mullipath discrimination, disadvantages exist in the form of higher equipment costs and lower flexibility in frequency assignment within the allocated bandwidth. In d spread spectrum cellular telephone system, such as illustrated in Figure 1, the preferred waveform design implemented involves a direct sequence pseudonoise spread spectrum carrier. The chip rate of the PN sequence is chosen to be 1.2288 MHz in the preferred embodiment. This particular chip rate is chosen so that the resulting bandwidth, about 1.25 MHz after filtering, is approximately one-tenth of the total bandwidth allocated to one cellular service carrier. •Another consideration in the choice of the exact chip rate is that it is desirable that the chip rate be exactly divisible by the baseband data rates to be used in the system. It is also desirable for the divisor to be a power of two. In the preferred embodiment, the baseband data rate is 9600 bits per second, leading to a choice of 1.2288 MHz, 128 times 9600 for the PN chip rate. In the cell-to-mobilc link, Ihc binary sequences used for spreading the spectrum are constructed from two different types of sequences, each with different properties to provide different functions. There is an outer code that is shared by all signals in a cell or sector that is used to discriminate between multipath signals. The outer code is also used to discriminate between signals transmitted by different cells or sectors to the mobile units. There is also an inner a^dc that is used to discriminate between user signals transmitted by single sector or cell. The carrier waveform design in the preferred embodiment for the cell-site transmitted signals utilizes a sinusoidal carrier that is quadraphase (four phase) modulated by a pair of binary PN sequences that provide the outer code transmitted by a single sector or cell. The sequences are generated by two different PN generators of the same sequence length. One sequence bi-phase modulates the in-phase channel (I Channel) of the carrier and the other sequence bi-phase modulates the quadrature phase (Q Channel) of the carrier. The resulting signals are summed to form a composite four-phase carrier. Although the values of a logical "zero" and a logical "one" arc conventionally used to represent the binary sequences, the signal voltages used in the modulation process arc +V volts for a logical "one" and -V volts for a logical "zero". To bi-phase modulate a sinusoidal signal, a zero volt average value sinusoid is multiplied by the +V or -V voltage level as controlled by the binary sequences using a multiplier circuit. The resulting signal may then be band limited by passing through a bandpass filter. Jt is also known in the art to lowpass filter the binary sequence stream prior to multiplying by the sinusoidal signal, thereby interchanging the order of the operations. A quadraphase modulator consists of two bi-phase modulators each driven by a different sequence and with the sinusoidal signals used in the bi-phase rtKxiulators having a 90° phase shift therebetween. In the preferred embodiment, the sequence length for the transmitted signal carrier is chosen to be 32768 chips. Sequences of this le-ngth can be generated by a modified maximal length linear sequence generator by adding a zero bit to a length 32767 chip sequence. The resulting sequence has good cross-correlation and autocorrelation properties. Good cross-correlation and autocorrelation properties are necessary to prevent mutual interference between pilot carriers transmitted by different cells. A sequence this short in length is desirable in order to minimize acquisition time of the mobile units when they first enter the system without knowledge of system timing. With unknown timing, the entire length of the sequence must be searched to determine the correct timing. The longer the sequence, the longer time the acquisition search will require. Although sequences shorter than 32768 could be used, it must be understood that as sequence length is reduced, the code processing gain is reduces. As processing gain is reduced, the rejection of multipath interference along with interference from adjacent cells and other sources will also be reduced, perhaps to unacceptable levels. Thus, there is a desire to use the longest sequence that can be acquired in a reasonable time. It is also desirable to use the same code polynomials in all cells so that the mobile unit, not knowing what cell it is in when initially acquiring synchronization, can obtain full synchronization by searching a single code polynomial. In order to simplify the synchronization process, all the cells in [he system „are synchronized to each other. In the exemplary 2mbodiment, cell synchronization is accomplished by synchronizing all [he cells to a common time reference, the Navstar Global Positioning System satellite navigation system which is itself synchronized to Universal Coordinated Time (UTC). Signals from different cells are differentiated by providing time affsets of the basic sequences. Each cell is assigned a different time offset o( the basic sequences differing fro^ its neighbors. In the preferred embodiment, the 32768 repetition period is divided into a set of 512 liming offsets. The 512 offsets are spaced 64 chips apart. Each sector of each cell in a cellular system is also assigned a different one of the offsets to use for all its transmissiorw. If thete are more than 512 sectors or cells in the system, then the offsets can be reused in the same manner as frequencies are reused in the present analog FM cellular system. In other designs, a different number than 512 offsets could be used. With reasonable care in assignment of pilot signal offsets, it should never be necessary for near neighboring cells to use near neighboring time offsets. All signals transmitted by a cell or one of the sectors of the cell share the same outer PN codes for the I and Q channels. The signals arc also spread with an inner orthogonal code generated by using Walsh functiorts. A signal addressed to a particular user is multiplied by the outer PN sequences and by a particular Walsh sequence, or sequence of Walsh sequences, assigned by the system controller for the duration of the user's telephone call. The same inner code is applied to both the I and Q channels resulting in a modulation which is effectively bi-phase for the inner code. it is well known in the art that a set of n orthogonal binary sequences, each of length n, for n any power of 2 can be constructed, see IJigital Communications with Space Applications, S.W. Golomb et al., Prentice-Hall, Inc, 1964, pp. 45-64. In fact, orthogonal binary sequence sets are also known for most lengths which are multiples of four and less than two hundred. One class of such sequences that is easy to generate is called the Walsh function, also known as Hadamard matrices. •. A Walsh function of order n can be defined recursively as follows: A Walsh sequence is one of the rows of a Walsh function matrix. A Walsh function of order n contains n sequences, each of length n bits. A Walsh function of order n (as well as other orthogonal functions) has the property that over the interval of n code symbols, the cross-correlation between all the di!fferent sequences within the set is zero, provided that the sequences ar^ time aligned with each other. This can be seen by noting that every isequence differs; from every other sequence in exactly half of its bits, it should also be noted that there is always one sequence containing all zeroes and that all the other sequences contain half ones and half zeroes. Neighboring cells and sectors can reuse the Walsh sequences because the outer PN codes used ih neighboring cells and sectors are distinct. Because of the differing propngation times for signals between a particular mobile's location and two or more different cells, it is not possible to satisfy the condition of time alignment required for Walsh function orthogonality for both cells at one time. Thus, reliance must be placed on the outer PN code to provide discrimination between signals arriving at the mobile unit from different cells. However, all the signals transmitted by a cell are orthogonal to each other and thus do not contribute interference to each other. This eliminates the majority of the interference in most locations, allowing a higher capacity to be obtained. The system further envisions the voice channel to be a variable rate channel whose data rate can be varied from data block to data block with a minimum of overhead required to control the data rate in use. The use of variable data rates reduces mutual interference by eliminating unnecessary transmissions when there is no useful speech to be transmitted. Algorithms are utilized within the vocoders for generating a varying number of bits in each vocoder block in accordance with variations in speech activity. During active speech, the vocoder may produce 20 msec, data blocks containing 20, 40, 80, or 160 bits, def)ending on the activity of the speaker. It is desired to transmit the data blocks in a fixed amount of time by varying the rate of transmission. Ii is further desirable not to require signalling bits to inform the receiver how many bits are being transmitted. The blocks are further encoded by the use of a cyclic redundancy check code (CRCC) which appends to the block an additional set of parity bits which can be used to determine whether or not the block of data has been decoded correctly. CRCC check codes are produced by dividing the data block by a predetermined binary polynomial. The CRCC consists of B11 or a pxjrticn of the remainder bits of the division process. The CRCC is checked in the receiver by reproducing the same remainder and checking to see of the received remainder bits are the same as the regenerated check bits. In the disclosed invention, the receiving decoder decodes the block as if it contains 160 bits, and then again as if it contains 80 bits, etc. until all possible block lengths have boon tried. The CRCC is computed for each trial decoding. If one of the trial decodings results in a correct CRCC, the data block is accepted and passed on to the vocoder for further processing. If no trial decoding produces a valid CRCC, the received symbols are passed on to the system's signal processor where other processing operations can optionally be performed. In the cell transmitter, the power of the transmitted waveform is varied as the data rate of the block is varied. The highest data rate uses the highest carrier power. When the data rate is lower than the maximum, the modulator, in addition to lowering the power, repeats each encoded data symbol a number of times as required to achieve the desired transmission rate. For example, at the lowest transmission rate, each encoded symbol is repeated four times. In the mobile transmitter, the peak p>ower is held constant but the transmitter is gated off 1/2, or 1/4 or 1/8 of the time in accordance with the number of bits to be transmitted in the data block. Tlie positions of the on-times of the transmitter is varied pseudo-randomly in accordance with the mobile user's addressed user code. Cell-to-Mobile Link In the preferred embodiment, the Walsh function size n, is set equal to sixty-four (n = 64) for the cell-to-mobile link. Therefore each of up to sixty-four different signals to be transmitted are assigned a unique orthogonal sequence. The forward error correction (FEC) encoded symbol stream for each voice conversation is multiplied by its assigned Walsh sequence. The Walsh coded/FEC encoded symbol stream for each voice channel is then multiplied by the outer PN coded waveform. The. resultant spread symbol streams are then added together to form a composite waveform. The resulting composite waveform is then modulated onto a sinusoidal carrier, bandpass filtered, translated to the desired operating frequency, amplified and radiated by the antenna system. Alternate embodiments of the present invention may interchange the order of some of the just described operations for forming the cell-site transmitted signal. For example, it may be preferred to multiply each voice channel by the outer PN coded waveform and perform the filtering operation prior to summation of all the channel signals to be radiated by the antenna. It is well known in the art that the order of linear operations may be interchanged to obtained various implementation advantages and different designs. The waveform design of the preferred embodiment for cellular service uses the pilot carrier approach for the cell-to-mobile link as described in Patent No. 4,901307. All cells transmit pilot carriers using the same 32768 length sequence, but with different timing offsets to prevent mutual interference. The pilot waveform uses the all-zero Walsh sequence, i.e. a Walsh sequence comprised of all zeroes that is found in all Walsh function sots. The use of the all-zero Walsh sequence for all cells' pilot carriers allows the initial search for the pilot waveform to ij',norc the Walsh functions until after the outer code PN synchronization has been obtained. The Walsh framing is locked to the PN code cycle by virtue of the length of the Walsh frame being a factor of the PN sequence length. Therefore, provided that the cell addressing offsets of the FN code are multiples of sixty-four chips (or the Walsh frame length) then the Walsh framing is known implicitly from the outer PN code timing cycle. All the cells in a service area are supplied with accurate synchronization. In the preferred errlbodiment, a GPS receiver at each cell synchronizes the local waveform timing to Universal Coordinated Time (UTC). The GPS system allows time synchronization to better than 1 microsecond accuracy. Accurate synchronization of cells is desirable in order to allow easy handoff of calls between cells when mobiles move from one cell to another with a call in progress. If the neighboring cells are synchronized, the mobile unit will not have difficulty synchronizing to the new cell thereby facilitating a smooth handoff. The pilot carrier is transmitted at a higher power level than a typical voice carrier so as to provide greater signal to noise and inlcrfcronce margin for this signal.The higher power level pilot carrier enables the initial acquisition search to be done at high speed and to niiiko possible a very accurate tracking of the carrier phase of the pilot carrier by a relatively wide bandwidth phase tracking circuit. The carrier phase obtained from tracking the pilot carrier is used as the carrier phase reference for demodulation of the carriers modulated by user information signals. This technique allows many user carriers to share the common pilot signal for carrier phase reference. For example, in a system transmitting a total of fifteen simultaneous voice carriers, the pilot carrier might be allocated a transmit power equal to four voice carriers. In addition to the pilot carrier, another carrier intended to be received by all system users in the eel! is transmitted by the cell-site. This carrier, called the synchronization channel, also uses the same 32768 length FN sequence for spectrum spreading but with a different, pre-assigned Walsh sequence. The synchronization channel transmits a broadcast message containing system information for use by the mobiles in the system. The system information identifies the cell-site and the system and conveys information allowing the long PN codes used for mobile information signals to be synchronized without addit4onal searching. Another channel, called the paging channel may be provided to transmit messages to mobiles indicating that a call ha^; arrived for them, and to respond with channel assignrncnls when a mobile initiates a call. Each voice carrier transmits a digital representation of the speech for a telephone call. The analog speech waveform is digitized using standard digital telephone techniques and then compressed using a vocoding process to a data rate of approximately 9600 bits per second. This data signal is then rale r ■= 1/2, constraint length K = ^i convolutional encoded, with repetition, and interleaved in order to provide error detection and correction fur\ctions which allow the system to operate at a much lower signal-to-noise and interference ratio. Techniques for convolutional encoding, repetition and interleaving arc weil known in the art. The resulting encoded symbols are multiplied by an assignee Walsh sequence and then multiplied by the outer PN code. This proces; results in a PN sequence rate of 1.2288 MHz or 128 times the 9600 bps dat; rate. The resulting signal is then modulated onto an RF carrier anc summed with the pilot and setup carriers, along with the other voici carriers. Summation may be accomplished at several different points in the processing such as at the IF frequency, or at the baseband frequency either before or after multiplication by the PN sequence. Each voice carrier is also multiplied by a value that sets its transmitted power relative to the power of the other voice carriers. This power control feature allows power to be allocated to those links that require higher power due to the intended recipient being in a relatively unfavoring location. Means are provided for the mobiles to report their received signal-to-noise ratio to allow the p>ower to be set at a level s Figure 2 illustrates in block diagram form an exemplary embodiment cell-site equipment. At the cell-site, two receiver systems are utilized with each having a separate antenna and analog receiver for space diversity reception. In each of the receiver systems the signals arc processed identically until the signals undergoes a diversity combination process. The elements within the dashed lines correspond to elements corresponding to the communications between the cell-site and one mobile unit. The output of the analog receivers are also provided to other elements used in communications with other mobile units. In Figure 2, the first receiver system is comprised of antenna 30, analog receiver 32, searcher receiver 34 and digital data receiver 36. The first receiver system may also include an optional digital data receiver receiver 38. The second receiver system includes antenna 40, analog receiver 42, searcher receiver 44 and digital data receiver 46. The cell-site also includes cell-site control processor 48. Control processor 48 is coupled to data receivers 36, 38, and 46 along with .searcher receivers 34 and 44. Control processor 48 provides among other functions, functions such as signal processing; timing signal generation; power control; and control over handoff, diversity, diversity combining and system control processor interface with the MTSO (Figure 8). Walsh sequence assignment along with transmitter and receiver assignment is also provided by control processor 48. Both receiver systems are coupled by data receivers 36, 38, and 46 to diversity combiner and decoder circuitry 50. Digital link 52 is coupled to receive the output of diversity combiner and decoder circuitry 50. Digital link 52 is also coupled to control processor 48, cell-site transmit modulator 54 and the MTSO digital switch. Digital link 52 is utilized to communicate signals to and from the MTSO (Figure 8) with cell-site transmit modulator 54 and circuitry 50 under the control of control processor 4S. The mobile unit Jransmilled signals are direct sequence spread spectrum signals that are modulated by a PN sequence clocked at a predetermined rate, which in the preferred embodiment is 1.2288 MHz. This clock.rate is chosen to be an integer multiple of the baseband data rate of 9.6 Kbps. Signals received on antenna 30 are provided to analog receiver 32. The details of receiver 32 are further illustrated in Figure 3. Signals received on antenna 30 are provided to downconverter 100 which is comprised of RF amplifier 102 and mixer 104. The received signals are provided as an input to RF amplifier where Ihey are amplified and output to an input to mixer 104. Mixer 104 is provided another input, that being the output from frequency synthesizer 106. The amplified RF signals are translated in mixer 104 to an IF frequency by mixing with the frequency synthesizer output signal. The IF signals are then output from mixer 104 to bandpass filter (BPF) 108, typically a Surface Acoustic Wave (SAW) filter having a passband of 1.25 MHz, where they are bandpass filtered. The filtered signals are output from BPF 108 to IF amplifier 110 where the signals are amplified. The amplified IF signals are output from IF amplifier 110 to analog to digital (A/D) converter 112 where they are digitized at a 9.8304 MHz clock rate which is exactly 8 times the PN chip rate. Although (A/D) converter 112 is illustrated as part of receiver 32, it could instead bo a part of the data and searcher receivers. The digitized IF signals are output from (A/D) converter 112 to data receiver 36, optional data receiver 38 and searcher receiver 34. The signals output from receiver 32 are I and Q channel signals as discussed later. Although as illustrated in 1-igure 3 with A/D converter 112 being a single device, with later splitting of the I and Q channel signals, it is envisioned that channel splitting may be done prior to digitizing with two separate A/D converters provided for digitizing the 1 and Q channels. Schemes for tho RF-'-IF-Baseband frequency downconvorsion and analog to eiigilal conversion for I and Q channels are Well known in the art. Searcher receiver 34 is used to at the cell-site to scan the time domain about the received signal to,ensure that the associated digital data receiver 36, and data receiver 38 if used, are tracking and processing the strongest available time domain signal. Searcher receiver 64 provides a signal to cell-site control processor 48 which provides control signals to digital data receivers 36 arid 38 for selecting the appropriate received signal for processing. The signal processing in the ciji'll-site data receivers and searcher receiver is different in several aspects than the signal processing by similar elementis in the mobile unit: In the inbound, i.e. reverse or mobile-tp-cell link, the mobile unit does not transmit a pilot signal that can be used for coherent reference purposes in signal processing at the cell-site. The mobile-to-cell link is characterized by a non-coherent modulation and demodulation scheme using 64-ary orthogonal signalling. In the 64-ary orthogonal signalling process, the mobile unit transmitted symbols are encoded into one of 2^, i.e. 64, different binary .sequences. TT»e set of sequences chosen are known as Walsh functions. The optimum receive function for the Walsh function m-ary signal encoding is the Fast Hadamard Transform (FHT). Referring again to Figure 2, searcher receiver 34 and digital data receivers 36 and 38, receive the signals output from analog receiver 32. In order to decode the spread spectrum signals transmitted to the particular cell-site receiver through which the mobile unit communicates, the proper PN sequences must be generated. Further details on the generation of the mobile unit signals are discussed later herein. As illustrated in Figure 3, receiver 36 includes tv^ro PN generators, PN generators 120 and 122, which generate two different short code PN sequences of the same length. These two PN sequences are common to those of all cell-site receivers and all mobile units with respect to the outer code of the modulation scheme as discussed in further detail later herein. PN generators 120 and 122 thus respectively provide the output sequences, PNj and PNQ. The PNj and PNQ sequences are respectively referred to as the In-Phase (I) and Quadrature (Q) channel PN sequences. The two PN sequences, PNj and PNQ, are generated by different polynomials of degree 15, augmented to produce sequences of length .^2768 rather than 32767 which would normally be produced. For example, the augmentation may appear in the form of the addition of a single zero to the run of fourteen O's in a row which appears one time in every maximal linear sequence of degree 15. In other words, one state of (he PN generator would be repeated in the generation o( the sequence Thus the modified Sequence contains one run of fifteen I's and t>ne run of fifteen O's. Such a PN generator circuit is disclosed in cof>ending U.S. Patent Application entitled "POWER OF TWO LENGTH PSEUDO-NOISE SEQUENCE GENERATOR. WITH FAST OFFSET ADJUSTMENTS", Serial No. 07/__^ , filed , and assigned to the assignee of the prescr^t invention. In the exemplary embodiment receiver 36 also includes a long code PN generator 124 which generates a PNy sequence corresponding to a PN sequence generated by the mobile unit in the mobile-to-cell link. PN generator 124 can be a maximal linear sequence generator that generates a user PN code that is very long, for example degree 42, time shifted in accordance with an additional factor such as the mobile unit address or user ID to provide discrimination among users. Thus the cell-site received signal is modulated by both the long code PN(j sequence and the short code PNj and PNQ sequences. In the alternative, a non¬linear encryption generator, such as an encryplor using the data encryption standard (DES) to encrypt a 64-symbol representation of universal time using a user specific key, may be utilized in place of PN generator 124. The PNy sequence output from PN generator 124 is exchisivc-OR'ed with the PNj and PNQ sequences respectively in exclusive-OR gates 126 and 128 to provide the sequences PNj' and PNQ'. The sequences PN[' and PNQ' are provided to PN QPSK correlator 130 along with the I and O channel signals output from receiver 32. Correlator 130 is is utilized to correlate the I and Q channel data with the Pr^l' and PNQ' sequences. The correlated I and QVhannel outputs of correlator 130 are respectively provided to accumulators 132 and 134 where the symbol data is accumulated over a 4-chip period. The outputs of accumulators 132 and 134 are provided as inputs to Fast Hadamard Transform (FHT) processor 136. FHT processor 146 produces a set of 64 coefficients for every 6 symbols. The 64 coefficients are then multiplied by a weighting function generated in control processor 48. The weighting function is linked to the demodulated signal strength. The weighted data output from I'HT 136 is provided to diversity combiner . and decoder ciricuity 50 (Figure 2) for further processing. The second receiver system processes the received signals in a manner similar to that discussed with resp>ect to the first receiver system of Figures 2 and 3. The weighted 64 symbols output from receivers 36 and 46 are provided to diversity combiner and decoder circuitry 40. Circuitry 50 includes an adder which adds the weighted 64 coefficients from receiver 36 to the weighted 64 coefficients from receiver 46. The resulting 64 coefficients arc compared with one another in order to determine the largest coefficient. The magnitude of the comparison result, together with the identity or the largest of the 64 coefficients, is used to determine a set of decoder weights and symbols for use within a Viterbi algorithm decoder implemented in circuitry 50. The Viterbi decoder contained within circuitry 50 is of a type capable of decoding data encoded at the mobile unit with a constraint length K = 9, and of a code rate r = 1/3. The Viterbi decoder is utilized to determine the most likely information bit sequence. Periodically, nominally 1.25 msec , a signal quality estimate is obtained and transmitted as a mobile unit power adjustment command along with data to the mobile unit. Further information on the generation of this quality estimate is discussed in further detail in the copending application mentioned above. This quality estimate is the average signal-lo-noise ratio over the 1.25 msec interval. Each data receiver tracks the timing of the received signal it is receiving. This is accomplished by the well known technique of correlating the received signal by a slightly early local reference PN and correlating the received signal with a slightly late local reference PN. The difference between these two correlations will average to zero if there is no timing error. ConverseIy,| if there is a timing error, then this difference will indicate the magnitude and sign of the error and the receiver's timing is adjusted accordingly. The cell-site further includes aluenna 62 which is couiiled to GPS i receiver 64. GPS receiver processes Signals received on antenna 62 from satellites in the Navstar Global Posijlioning System satellite navigation system so as to provide timing signals indicative of Universal Coordinated Time (UTC). GPS receiver 64 provides these timing signals to control processor 48 for timing; synchronizing at the cell-site as discussed previously. ' Circuitry 50 includes an adder which adds the weighted 64 coefficients from receiver 36 to the weighted 64 coefficients from receiver 46. The resulting 64 coefficients are compared with one another in order to determine the largest coefficient. The magnitude of the comparison result, together with the identity or the largest of the 64 coefficients, is used to determine a set of decoder weights and symbols for use within a Viterbi algorithm decoder implemented in circuitry 50. The Viterbi decoder contained within circuitry 50 is of a type capable of decoding data encoded at the mobile unit with a constraint length K = 9, and of a code rate r = 1/3. The Viterbi decoder is utilized to determine the most likely information bit sequence. Periodically, nominally 1.25 msec , a signal quality estimate is obtained and transmitted as a mobile unit power adjustment command along with data to the mobile unit. Further information on the generation of this quality estimate is discussed in further detail in the copending application mentioned above. This quality estimate is the average signal-to-noisc ratio over the 1.25 msec interval. Each data receiver tracks the jtiming of the received signal it is receiving. This is accomplished by the well known technique of correlating the received signal" by a slightly early local reference PN and correlating the received signal with,a slightly late local reference PN. The difference between these two correlations will average to zero if there is no timing error. Conversely^ if there is a timing error, then this difference will indicate the magnitijde and sign of the error and the receiver's timing is adjusted accordinHy. The cell-site further includes antenna 62 which is coupled to GF'S receiver 64. GPS receiver processes signals received on antenna 62 from satellites in the Navstar Global Posiitioning System satellite navigation system so as, to .pxovide timing | signals indicative of Universal Coordinated Tinie (UTC). GPS receiver 64 provides these timing signals to control processor 48 for timing; synchronizing at the cell-site as discussed previously. In Figure 2 optional digital data receiver 38 may be included for improved performance of the system.The structure and operation of this receiver is similar to that described with reference to the data receivers 36 and 46. Receiver 38 may be utilized at the cell-site to obtain additional diversity modes. This additional data receiver alone or in combination with additional receivers can track and receive other possible delay paths of mobile unit transmitted signals.. Optional additional digital data receivers such as receiver 38 provides additional diversity modes which are extremely useful in those cell-sites which are located in dense urban areas where many possibilities for muUipath signals occur. Signals from the MTSO are coupled to the appropriate transmit modulator via digital link 52 under control of control proco.s.sor 48. Transmit modulator 54 under control of control processor 48 spread spectrum modulates the data for transmission to the intended recipient mobile unit. Further details with respect to the structure and operation of transmit mcxiulator 54 are di.scussed Lx'Iow with reference to Figure 4. The output of transmit modjulator 54 is provided to transmit !■ power control circuitry 56 where unjder the control of control processor 48 the transmission power may be controlled. The output of circuitry 56 is provided to summer 57 where lit is summed with the output of transmit modulator/transmit power control circuits directed to other mobiles in the cell. The output of .Rummer 57 is provided to transmit power amplifier circuitry 58 where q'utput to antenna 60 for radiating to mobile units within the cell service area. Figure 2 further illustrates pilot/control channel generators and; transmit power control circuitry 66. Circuitry 66 under control of conlifol processor generates and power controls the pitot signal, the sync channel, and the paging chaimel for coupling to drcuitry 58 and output to antenna 60. A block diagram of an exenjiplary embodiment, of the ceU-site transmitter is illustrated in Figure 4i The transmitter includes a pair of PN sequence generators used in generating the outer code. These PN generators generate two different I'JSI sequences, i.e. the I'Nf and PNQ sequences, as was discussed with reference to Figure 3. I lowcver, these PNi and PNQ sequences are delayed in time according to the sector or cell ndd ress. In Figure 4, the transmitter circuitry of Figure 3 is illustrated in further detail with the pilot, sync, paging and voice channel signals. The transmitter circuitry includes two FN generators, FN generators 196 and 198, which generate the PNj and PNQ sequences. PN generators 196 and 198 are responsive to an input signal corresponding to a sector or cell address signal from the control processor so as to provide a predetermined time delay to the PN sequences. These time delayed PNi and FNQ sequences again relate respectively to the In-Phase (I) and Quadrature (Q) channels. Although only two PN generators are illustrated for respectively generating the PNj and PNQ sequences for the corresponding channels of the cell-site or sector, it should be understood \hi\[ many other PN generator sclu'int^s may be itnplemonled. For (^xample, in a un.sectorized cell, a pair of PN generators may be provided for each of the pilot, sync, paging and voice channels to produce, in synchronization, the PNj and PNQ sequences used in the outer code. Such, a case may be advantageous to avoid distributing the PNi and PNQ scciuenccs throughout a large number of circuits. In the preferred embodiment> Walsh function encoding of the channel signals is employed as th,^ inner code. In the exemplary numerology as disclosed herein, a tojtal of 64 different Walsh sequences are available with three of these sequences dedicated to the pilot, sync and paging channel functions. In the sync, paging and voice channels, input data is convolutionally onccHlJHl and then interleaved as is well known in the art. Furlhormore, the convolulional encoded data is al.so provided with repetition before interleaving as is also well known in the art. I The pilot channel contains no data modulation and is chafacterized as an unmodulated spread spectrum signal that all of the users of a particular cell-site or sector use for acquisition or tracking purposes. Each cell, or if divided injto sectors, each sector has a unique pilot signal. However, rather than using different PN generators for the pilot signals, it is realized that a more efficient way to generate different pilot signals is to use shifts in the same basic sequence. Utilizing this technitjue a mobile unit sequentially searches the whole sequence and tunes to the offset or shift that produces the strongest correlation. In using this shift of the basic sequence, the shifts must be such that the pilots in adjacent cells or sectors must not interfere or cancel. The pilot sequence must thereft)re bo long enough that many different sequences can be generated by .-ihifts in the basic .sequence to support a large number of pilot signals in the system. Furthermore, the separation or shifts must be great enough to ensure that there is no interference in pilot signals. Accordingly, in a exemplary embodiment of the present invention the pilot sequence length is chosen to IH» 2^'. The sequence is generated started by a sequence 2^^-1 with an extra 0 appended to the sequence when a particular slate is delected. In the exemplary embodiment there are chosen to be 512 different pilot signals with off.sets in the basic sequence of 64 chips. However, offsets may be integer multiples of the 64 chip offset iwith a corresponding reduction in the number of different pilot signals. ' In generating lh With respect to the portion df transmit modulator and power control circuitry 66 for Ihe pilot channel, Walsh geni'raU)r (W()) 2(){) generates a signal corresponding :o the all zero function as just discussed. The timing in the generation of the Walsh function is provided by the control proces.sor, a.i in the ca.se of all Walsh fiuuiion generators in the cell-site and mobilejunit. The output of generator 200 is provided as an .nput to both of excIusive-OK gales 202 and 2{M. The other input of exclusivc-OR gale 202 receives the PN| signal wliile the other input of exclusive-OR gate 204 receives the PNQ signal. The PNi and PNQ signals are respectively exclusive-OR'ed with the output of generator 200 and respectively provided as inputs to Finite Impulse Response (I'lR) filters 206 and 208. The filtered signals output from I*"IR filters 206 and 208 provided to a transmit power control circuitry comprised of gain control elements 210 and 212. The signals provided to gain control elements 210 and 212 are gain controlled in response to input signals (not shown) from the control processor. The signals output from gain control elements are provided to transmit power amplifier circuitry 58 whose detailed structure and function is described later herein. The sync channel information is encoded and then multiplied in exc!usive-OR gates by a preassigned Walsh sequence. In the exemplary embodiment, the selected Walsh function is the (W32) sequence which consists of a sequence of 32 "ones" followed by 32 "zeros". The resulting sequence is then multiplied by the PNi and PNQ sequences in exclusive-OR gates. In the exemplary embodiment the sync channel dala informali. a is provided to the lransi;nit modulator typically at a rate of 1200 bps. In the exemplary embodiment the sync channel data is preferably convolutionally encoded ^t a rate r = 1/2 with a constraint length K - 9, with each code symbol;repeated twice. This encoding rale and constraint length is common to jail encoded forward link channels, i.e. sync, paging and voice. In an exemplary embodiment, a shift register .structure is employed for the generators of the ctxle G] - 753 (octal) ami C^2 ; ^A (octal). The symlx)! rale to he sync ch.umel is in Ihe x*'inplary embodiment 4800 sps, i.e. one symboli is 208 jisec or 256 PN chips. The code symbols are inlerleaved t)y means ol a convt>lulional inlerleaver spanning in iho ox thai any two symbols in a contiguous sequence of I-l or fewer symbols are separated by at least J + 1 symbols in a deinterlcaver output. r(]uivalenlly, any two symbols in a contiguous sequence of ]-l symbols are separated by at least I+l symbols at the deinterleaver output. In other words, if I = 16 and J = 48, in a string of 15 symbols, the symbols are transmitted separated by 885 psec, thus providing time diversity. The sync channel symbols of a particular cell or sector are tied to the corresponding pilot signal for thai cell or sector. Figure 5 illustrates the timing of two different pilot channels (N) and (N + 1) which arc separated by a shift of 64 chips, l-igure 5 illustrates only by way of example a timing diagram for the exemplary pilot and sync channels with the state of the actual pilot signal chips and sync channel symbols not illustrated. Each sync channel starts a new interleaver cycle with the first code symbol (cx) of a code symbol pair (cx, c'x), due to a code repeat of two, shifted with respect to absolute; time by an amount eijual to the corresj^xinding pilot. As illustrated in Figure 5, Tjie N pilot channel starts a new interleaver cycle, or pilot sync, at th^ time tx- Similarly, the N+1 pilot channel starts a new interleaver cycle or pilot sync at the time ty which occurs 64 chip later in time than time tx- l^e pijot cycle in the exemplary embodiment is 26.67 msec long, whi([:h corresponds to 128 sync channel code symbols or 32 sync channel information bits. The sync channel symbols are interleaved by a convolutional interleaver which spans 26,67 msec. Thus, when the mobile i nit has acquired the pilot signal, it has immediate sync channel interleai^er synchronization. The sync channel symbols are covered by the preassigned Walsh sequence to provide orthogonality iilv the signal. In the sync channel, one code symbol spans four cover seq'uences, i.e. one code symbol to four repetitions of the "32 one"-"32 /JCTO" .sequence, as illustrated in I'igure 6. As illustrated in Figure 6, a single logical "one" represents the occurance of 32 "one" Walsh chips while a single logical "zero " represents the occurance of 32 "zero" Walsh chips, (prlhogonalily in the sync channel is still maintained even though the sync channel symbols arc skewed with respect to absolute time depending upon the associated pilot channel because sync channel shifts are integer multiples of the Walsh frame. The sync channel messages in the exemplary embodiment are variable in length. The length of the message is an integer multiple of 80 msec which corresponds to 3 pilot cycles. Included with the sync channel information bits are cyclic redundancy (CRC) bits for error detection. Figure 7 illustrates in the form of a liming diagram the overall exemplary system timing. In the period of two seconds there are 75 pilot cycles. In Figure 7, the N pilot and sync channels correspond to the .sector or cell using the unshifted pilot such that the pilot and sync signals align exactly with UTC time. As such the pilot sync, i.e. initial .slale, aligns exactly with a common 1 pulse per second (pp.s) sij»,nal. In all cases in which a shifted pilot is used, a PN phase offset corre.sponding to the pilot .shift is intrbduced. In other words, pilot sync (initial state) and sync channel messages are skewed with respect to the 1 pps signals. The sync messages carries this pha.se off.set information so lli.ll llie mobile unit can adjusts its liiuing accordingly. As soon as a sync channel mes$age has been correctly received, the mobile unit has the ability lo imrncditUciy syiu hruni/x* lo oilluT a paj»in)', channel or a voice channel. At pilot syn'c, corresp>onding to the end of each sync message, a new 40 msec injterleaver cycle begins. At that time, the mobile unit starts deinterleaving the first code symbol of either a code repetition, or a (cx, Cx+i) pair, with decoder synchronization achieved. The deinlerleaver write adjdress is initialized to 0 and the read address is initialized to J, memor| deintcrleaver synchronization is .ichjoved. The sync channel messages cany information regarding the state of a 42-l)it long I'N gtMU>ralt)r for I he voice ihannt>l a.ssij'.ned for iht-communication with the mobile unit; This information is used at the mobile unit digital data receivers to! synchronize the corresponding PN );enerators. For example, in Figur^ 7 the sync channel message N + 1 contains a 42-bit field which is indicative of the slate, slate X, that the sector or cell voice channel corresponding long code PN generator will hnvc at a predetermined later time, such as 160 msec later. The mobile unit, after successfully decoding a sync channel message, loads at the correct instant of time the long ccxie TN generator with the state X. The mobile unit long code PN generator is thus synchronized to permit (i(^scrainbling of the ust^r intended nu>ss Walsh generator 218 generates a signal ct)rresponding to the Walsli (W32) .sequence that is provided as the other input to exclusive-OR gale 216. The sync channel syrtibol stream and the Walsh (W32) si'ijuence are excIusive-OR'ed by exc;lusive-C)K g,ale 216 wiili (he result thereof provided as an input to both of exclusive-OR gates 220 and 222. The other input of exclusive-QR gate 220 receives the PNj signal while the olher input of exclusive-OK gate 222 receives the I'NQ signal. The PNi and PNQ signals are respectively exclusive-OR'ed with the output of exclusive-OR gate 218 and! respectively provided as inputs to Finite Impulse Response (FIR) fillers 224 and 226. The filtered signals i)utput from l resulting sequence is then multiplied by the I'Nj and I'NQ sequences. The data rate of the paging channel for a particular sector or cell is indicated in an assigned field in the sync channel message. Although the paging channel data rate is variable, it is in the exemplary embodiment fixed for each system at one of the following exemplary data rales: 9.6, 4.8, 2.4 and 1.2 kbps. With respect to the transmit modulator and power control circuitry of the paging channel, the paging channel information is input from the control processor to encoder 232. Encoder 232 is in the exemplary embodiment a convolutional encoder that also provides repetition of the symbols according to the assigned data rate of the channel. The output of encoder 232 is provided to interleaver 233 where the symbols are convolutional interleaved. The output from interleaver 233 is provided as an input to exclusivc-OR gate 234. Although the paging channel data rate will vary, the code symbol rale is kept constant at 19.2 ksps by CCK1(> rep>elilion. Walsh generator 236 generates a signal, corresponding to a preassigned Walsh sequence, that is provided as the othci input .to i'xcliisiv(>-()R gal«" 2.34. 'I'hc symbol d.il.i ,ind Walsh S The data of each voice channel is also encoded with repetition, interleaved, scrambled, multiplied by its assigned Walsh sequence (Wi -Wj), and then multiplied by the rN| and TNQ sequences. The Walsh sequence to be used by a particular channel is assigned by the system controller at call setup time in the same manner as channels arc assigned to calls in the analog FM cellular system. In the exemplary embodiment illustrated herein, up to 61 different Walsh sequences arc available for use by the voice channels. In the exemplary embodiment of the present invention, the voice channel utilizes a variable data rate. The intent in using a variable data rate is to lower the data rate when there is no voice activity thereby reducing interference generated by this particular voice chani^el to other users. The vocoder envisioned to provide variable rate data is disclosed in copending U.S. Patent Application "VARIABI.F. IMTF, VCX^ODF.R" Serial No. , filed _ _, also assigned to the a.ssignee of the present invention. Such a vocoder produces data at four different data rates based on voice activity on a 20 m.sec frame basis. Exemplary data rates are 9.6 kbps, 4.8 kbps, 2.4 kbps and 1.2 kbps. Although the data rate will vary on a 20 m.sec basi.s, the code symlx>l rate is kept constant by code repetition at 19.2 ksps. Accordingly, the code symbols are repeated 2, 4 and 8 times for the respective data rates 4.8 kbp>s, 2.4 kbps and 1.2 kbps. Since the variable rate scheme is devised to reduce interference, the code symbols at the lower rates will have lower energy. For example, for the exemplary data rates of 9.6 kbps, 4.8 kbps, 2.4 kbps and 1.2 kbps, the code symbol energy (Fs) is respectively F.b/2, Fi,/4, I'b/8 and F\i,/16 where E\) is the information bit energy for ihe 9.6 kbps transmission rate. The code symbols are inlerleiived by a convolutional interleaver such that rode symbols with differerit energy levels ^vill be scrambled by the operation of the rnlerleaver. Inlbrderlo keep track of what energy level a code symbol should have a label is attached to each symbol specifying its data rate for scaling [purposes. After orthogonal Walsh covering and PN spreading, the quadrature channels are digitally filtered by a Finite Impulse Response (FIR) filler. Tlie FIR filler will receive a signal corresponding to the symbol energy level in order to accomplish energy scaling according to the data rale. The I and Q channels will be .scaled by factors of: 1,1/^2, 1/2, or 1/2V2 . In one implementation the vocoder would provide a data rate lalx'l in the form of a 2-bit number to the h'lR filter for controlling the filter scaling coefficient. In Figure 4, the circuitry of (wo exemplary voice channels, voice channels (i) and (j) are illustrated. The voice channel (i) data is input from an associated vocoder (not shown) to transmit modulator 54 (I'igure 3). Transmit modulator 54 is comprised of encoder 250i; interleaver 251i; exclusive-OR gates 252i, 255i, 256i and 258;; PN generator 253i; and Walsh generator (W;) 254i. The voice channel (i) data is input to encoder 250; where in the exemplary embodiment it is convolutional encoded with code symbol repetition according to the input data rale. The encoded data is then provided to interleaver 251 j where, in the exemplary embodiment, it is convolutional interleaved. Interleaver 251 j also receives from the vocoder associated with the voice channel (i) a 2-bit data rate label ttiat is mterleaved with the symbol data to identify at the data rate to the FIR filters. The data rate label is not transmitted. At the mobile unit, the decoder checks for all possible codes. The interleaved symbol data is output from interleaver 251i at an exemplary rate of 19.2 ksps to an input of exclusive-GR gate 252i. In the exemplary embodiment, each voice channel signal is scrambled to provide greater security in cell-to-mobile transmissions. Although such scrambling is not required it d(X>s enhance the .security in conununications. For example, scrainbiing of the voice channel signals may be acc(^mplished by PN cotiing the voice channel signals with a PN code determined by the mobile unit address of user II). Such scrarhbling may use the PNjj sequence or encryption scheme^as discussed with reference to Figure 3 with respect to the particular receiver for the mobile-to-ccll communications. Accordingly, a separate PN generator may be implemented for this function as illustrated in Figure 4. Although scrambling is discussed with reference to a PN sequence, scrambling may be accomplished by other techniques including those well known in the art. Referring again to Figure 4, scrambling of the voice channel (i) signal may be accomplished by providing PN generator 253i which receives the assigned mobile unit address from the control processor. PN generator 253i generates a unique PN code that is provided as the other input to exclusive-OR gate 252}. The output of exchisive-OR gate 252j is instead provided to the one input of excIusive-OR gate 255i. Walsh generator (Wj) 254] generates, in response to a function select signal and timing signals from the control processor, a signal corresponding to a preassigned Walsh sequence. The value of the function select signal may be determined by the address of the mobile unit. The Walsh sequence signal is provided as the other input tc exclusive-OR gate 255i. The scrambled symbol data and Walsh sequence are exclusive-OR'ed by exclusive-OR gate 255j with the result provided as an input to both of exclusive-OR gates 256; and 258i. PN generatoi 253i along with all other PN generators and Walsh generators at the cell-site provide an output at 1.22H8 MHz. it should be noleii that I'N generator 253 includes a decimator which provides' an output at a 19.1 kHz rate to exclusive-OR gate 255,. The other input of exclusive-OR gate 256^ receives the PNj signa while the other input of exclusive-OR gate 258i receives the PNQ signal The PN| and PNQ signals are respectively exclusive-OR'ed with lh output from gain control elements are provided to transmit power amplifier circuitry 58. In adiiilion to voice bits, the forw.ird link voice cliannel carries power control information. The power control bit rate is in the exemplary embodiment 800 bps. The cell-site receiver which is demodulating the mobile-to-cell signal from a given mobile, generates the power control information which is inserted in the cell-to-mobile voice channel addressed to that particular mobile. Further details on the power control feature is disclosed in the above identified copending application. Power control bits are inserted at the output of the convolutional intcrlcaver by means of a technique called code symbol puncturing. In other words, whenever a power control bit needs to be transmitted two code symbols are replaced by two identical code symbols with polarity given by the power control information. Moreover, power control bits are transmitted at the energy level corresponding to the 9600 bps bit rate. An additional constraint imposed on the power control information stream is that the position of the bits must be randomized among mobile-to-cell channels. Otherwi.se the full energy power control bits' would generate spikes of interference at regular intervals, thus diminishing the detectability of such bits. Figure 4 further illustrates voice channel (j) which is identical in function and structure to that of voice channel (i). It is contemplated that there exist many more voice chnnntls (not illustrated) with the total of voice channel being up to 61 for the illustrated embodiment. With respect to the Walsh generators of Figure 4, Walsh functions are a set of orthogonal binary sequSences that can be easily generated by means well knpwn in the art. Tliej characteristic of interest in the Walsh , function is that eacn of the 64 soqiiences is perfectly orthogonal to all of the other sequences. As such, any! pair of sequences differ in exactly as many bit positions as they agree, i.e. 32 over an interval of 64 symbols, rivus when information is encoi^ed for transmission by the Walsh sequences the receiver will be able to select any one of the Walsh sequences as a desired "carrier" signal. Any signal energy encoded onto the other Walsh sequences will be rejected and not result in mutual interference to the desired one Walsh sequence. In the exemplary embodiment for the cell-to-mobile link, the sync, paging and voice chanr\els as mentioned previously use convolutional encoding of a constraint length K = 9 and code rate r = 1/2, that is, two enccxicd symbols are produced and transmitted for every information bit to be transmitted. In addition to the convolutional encoding, convolutional interleaving of symbol data is further employed. U is further envisioned that repetition is also utilized in conjunction with the convolutional encoding. At the mobile unit the optimum decoder for this type of code is the soft decision Viterbi algorithm decoder. A standard design can be used for decoding purposes. The resulting decoded information bits are passed to the mobile unit digital baseband I'cjuipment. Referring again to Figure 4, circuitry 58 includes series of digital to analog ( D/A) converters for converting the digital information from the I'Ni and TNQ spread data for the pilot, sync, paging and voice channels to analog form. In particular the pilot-channel PN{ spread data is output froipn gain control element 210 lu D/A converter 268. The digili/.cd data is output from D/A converter 268 to an summer 284. Similarly, the output of the corresponding gain control elements for the sync, paging and voice channels PNi spread data, i.e. gain control elements 228, 246, and 264i - 264j, are resp)€Ctively provided to D/A converters 272, 276 and 280i - 280j where the signals are diratized and provided to summer 284. The PNQ spread data for the pilot, sync, paging and voice channels are -output from g^in control element.'^ 221, 230, 248, and 266^ - 266j, are respectively provided to D/A converters 270, 274, 278 and 282i - 2a2j where the signals are digitized and provided to sunwner 286. I Summer 284 sums the PNj spread data for the pilot, sync, paging and voice channels and while summer 286 sums the and PNQ spread data,for the same channels. The summed I and Q channel data is respectively irtput along with loc^l oscillator (LO) frequency signals Sin(2nfl) and Cos(2Kft) to mixers 288 and 290 where they are mixed and provided to summer 292. The LO frequency signals Sin(27ift) and Co,s(2Kfl) are provided from suitable frecjuency sources (not shown). . These mixed IF signals are summed in summer 292 and provided to mixer 294. Mixer 294 mixes the summed signal with an RF frequency signal provided by frequency synthesizer 296 so as to provide frequency upconvcrsion to the RF frequency band. The RF signal output from mixer 294 is bandpass filtered by bandpass filter 298 and output to RF amplifier 299. Amplifier 299 amplifies the band limited signal in accordance with the input gain control signal from the transmit power ronlrol circuitry ."6 (Figure 3). It should be understood that llio embodiment illustrated for transmit power amplifier circuitry 58 is merely for purposes of illustration with many variations in signal summing, mixing, filtering and amplification possible as is well known in tlie art. Cell-site control processor 48 (lMj;iire 3) has the responsibility for assignment of digital data receivers and transmit modulators to a p.irlicular call. Control processor 48 also monitors the proj'.ress of the call, quality of the signals and initiates teardown on loss of signal. The cell-site communicates with the MTSO via link 52 where it is coupled by a standard telephone wire, optical filx'r, or microwave link. Figure 8 illustrates in block diagram form the equipment utilized in the MTSO. The MTSO typically includes a system controller or control processor 300, digital switch 302, diversity combiner 304, digital vocoder 306 and digital switch 308. Although not illustrated additional diversity combiners and digital vocoders are coupled between digital swil(hes302 and 308. When the cell-diversity mcxle is active, the call is prt)ces.sed by two cell-sites. Accordingly, signals will arrive at the MTSO from more than one cell-site with nominally the same information. However, because of fading and interference on the inbound or reverse link from the mobile unit to the cell-sites, the signal from one cell-site may be of better qunlily than the signal from the other cell-site. Digital switch 302 is used in routing the information stream corresponding to a given mobile unit from one or more cell-sites to diversity combiner 304 or the corresponding diversity combiner as determined by a signal from system control processor 300. When the system is not in the cell diversity mode, diversity combiner 304 may be either bypassed or fed the same information on each input p>orl. A multiplicity of serial coupled diversity combiners and vocoder are provided in parallel, nominally one for each call to be processed. Diversity combiner 304 compares the signal quality indicators accompanying the information bits from the two or more cell-site signals. Diversity combiner 304 selects the bits corresponding to the highest quality cell-site on a frame-by-frame basis of the information for output to vocoder 306. Vocoder 306 converts the format of the digitized voice signal to standard 64 Kbps PCM telephone formal, analog, or any other standard format. The resultant signals is transmitted from vocoder 306 to digital switcl. 308. Under the control of system control processor 300, the call is routed to the PSTN. j Voice signals coming from the PSTN intended for the mobile units, are provided to digital switch 308 ft)r coupling ti) an appropriate I digital vocoder such as vocoder 306 under control of system control prcK'essor 300. Vocoder 306 encodesjthe input digitized voice signals and provides the resulting information fcil stream directly to digital switch 302. Digital switch 302 under systerp control processor control direct the encoded data to the cell-site or ce'll-siles to which the mobile unit is communicating. Althou'gli tliscijssed previously that inft)rmation transmitted to the MTSO analog jvoice, it lis further envisioned that digital information may also be communicated in the system. To ensure compatibility with the system, care niusl Ix' laktMi in proper framing of the data. If the mobile unit is in a handoff mode communicating to multiple cell-sites or in a cell diversity mode, digital switch 302 routes Iho calls to the appropriate coll-silos for transmission by the appropriate cell-site transmitter to the intended recipient mobile unit. However, if the mobile unit is communicating with only a single cell-site or not in a cell diversity mode, the signal is directed only to a single cell-site. System control processor 300 provides control over digital switches 302 and 306 for routing data to and from the MTSO. System control processor 300 also determines the assignment of calls to the cell-sites and to the vocoders at the MTSO. Furthermore, system control processor 300 communicates with each cell-site control processor about the assignment of particular calls between the MTSO and cell-site, and the assignment of PN codes for the calls. It should be further understood that as illustrated in Figure 8 digital switches 302 and 306 are illustrated as two separate switches, however, this function may be performed by a single physical switching unit. When the cell-diversity mode is in use, the mobile unit will use the searcher receiver to identify and acquire the strongest multipath signal from each of the two cell-sites. The digital data receivers will be controlled by the searcher receiver and the control processor so as to demodulate the strongest signals. When the number of receivers is less than the number of cell-sites transmitting information in parallel, a switching diversity capability is possible. For example, with only a single data receiver and with two cell-sites transmitting, the searcher will monitor the pilots from both cell-sites and choose the strongest signal for the receiver to demodulate. In this etnlx^diment the choice can be made as frequently as every vocoder frame, :or about every 20 msec. I The system control processor has responsibility for assignment of i- digital data receivers and modulators at the cell-site to handle particular calls. Thus in the cell-to-mobile link, the system control processor controls the assignment of Walsh sequences used at the cell-site in transmission of a particular call to j the mobile unit. In addition the system control proces.sor controls th(f receiver Walsh sequences and PN codes. In the mobilo-lo-coll link, the system control processor also controls the mobile unit user PN codes for the call. Assignment information is therefore transmitted from the MTSO to the cell-site and from there to the cell to the mobile. The system control processor also monitors the progress of the call, the quality of signals, and initiates tear down on loss of signal. Mobile-to Cell Link In the mobile-to-cell link, the channel characteristics dictate that the modulation technique be modified. In particular, the use of a pilot carrier as is used in the cell-to-mobile link is no longer feasible. The pilot carrier must be more powerful than a voice carrier in order to provide a good phase reference for data modulation. With the cell-site transmitting many simultaneous voice carriers, a single pilot signal can \x; shared by all the voice carriers. Therefore, the pilot signal power per voice carrier is quite small. In the mobile-lo-ccil link, however, there is usually only a single voice carrier per mobile. If a pilot were used, it would require significanlly more power than the voice carrier. This situation is cK jrly " not desirable since overall system capacity would be greatly reduced due to the interference caused by the presence of a larger number of high power pilot signals. Therefore, a modulation capable of efficient demodulation without a pilot signal must be used. With the mobile-to-cell channel corrupted by Rayleigh fading, resulting in a rapidly varying channel phase, coherent demodulator technicjues, such as a Coslas l(X>p which derives ph.ise from the receivetl signal, arc not feasible. Other techniques such as differentially coherent PSK can be employed but fail to prc)vide the desired level oi signal-lo-noise ratio performance. Thus, a form of orthogonal signaling such as binary, quaternary or m-ary signalling sn;juld be-empIoyea> In the exemplary embodiment, a 64-ary orthogonal signaling technique is employed using Walsh functions. The demodulator for mlarv orthom)nal sijinalinj; rec|uires channel coherence only over Ihc duration of transmission of the m-ary symbol. In the exemplary embodiment, this is only two bit times. The message encoding and modulation process begins with a convolutional encoder of constraint length K = 9 and code rate r = 1/3. At a nominal data rate of 9600 bits per second, the encoder produces 28800 binary symbols per second. These are grouped into characters containing 6 symbols each at a rate of 4800 characters per second with there being 64 possible characters. F.ach character is encoded into a length 64 Walsh sequence containing 64 binary bits or "chips." The 64-ary Walsh chip rate is 307,200 chips per second in the exemplary embodiment. The Walsh chips are then "ct>vered" or multiplied by a FN sequence running at the rate of 1.22K8 Ml Iz. i-lach mobile unit is assignee! a unique PN sequence for this purpose. This PN sequence can either be a.ssigned only for the duration of the call or assigned permanently to the mobile unit. The assigned PN sequence is referred to herein as the user PN sequence. The user PN sequence g.eneralor runs at a clock rate of 1.2288 MHz and so as to produce fouriPN chips for every Walsh chip. Finally, a pair of short, length 3?7(-,H, PN .sequences are generated. in the exemplary embodiment, the samt» sequences arc used as for the cell-to-mobile link. Tlie user PN sequence covered Walsh chip sequence is then covered or multiplied by each of the two short PN sequences. The two resulting sequence., then bi-phase modulate a quadrature pair of sinusoids and are summed into a single signal. The resulting signal is i then bandpass filtered, translated to the final RF frequency, amplified, filtered and radiated by the antenna of the mobile unit. As was discussed with reference to the cell-to-mobile .signal, the ordering of the filtering, i .implification, Iranslalion and modulation operations may be interchanged. ) In an alternative eml'H)diment, itwo different pha.ses of the user PN code might h>e produced and u.sed to"modulaU' the two carrier pha.ses of the quadraphase waveform, ilispensing with the need for using the length 32768 sequences. In yet another alternative, the mobile-to-cell link might utilize only bi-phnsc modulation, also dispensing with the need for the short sequences. The cell-site receiver for each signal produces the short PN sequences and the Mser PN sequence for each active mobile signal being received. TTie receiver correlates the received signal energy with each of the coded waveforms in separate correlators. Hach of the correlator outputs is then separately processed to demodulate the M-ary encoding and the convolufional coding using a Fast Jfadamard Transform processor and a Viterbi algorithm decoder. In another alternative modulation scheme for the mobilc-to-cell link, the same modulation scheme would be used as for the cell-to-mobile link. Each mobile would ulili/.e the pair of 32768 length sector codes as outer codes. The inner code would utilize a length 64 Walsh sequence that is assigned lo the mobile for u.se while it is in that sector. Nominally, the same Walsh sequence would be assigned to the mobile for the mobile-to-cell link as is used for the cell-lo-mobile link. The above orthogonal PN coding, scheme limits the available bandwidth spreading that can be used by the modulation system to a maximum rate of the chip rale diviiled by 6-1, or 1*.>20() I I/, for the numbers used in the exemplary embodiment. Tliis would preclude the use of m-ary encoding with large m as described for the exemplary embodiment. As an alternative, however, a rate r = 1/2, constraint length K = 9 convolutional code could be used with differential binary phase shift keying modulation of the encoded binary symbols. The demodulator in the cell-site could build up a pha.se reference over a siiorl interval using the technique! described in the ,»rticle "Nonlinear Hslimation of I*SK-ModuIafed Carrier with Application to Burst Digital Transmission", Andrew j. Vilerbi and Audrey M. Viterbi, UlliU Transactions On Information Theoify, Vol IT-29, No. 4, July 1983. For example, a phase reference could, be averaged over only 4 symbols requiring no more channel coherence than the alx)ve 64-ary scheme. The performance of the just described alternative scheme, however, will be inferior to the preferred embcxliment in Iho presence of severe Raylcigh fading and mullipath conditions. However, in certain environments where fading and mullipath are less severe, for example, the satellite-mobile channel and in certain land-mobile channels, the performance of the alternative system could be better than the preferred embodiment. This can occur Ix'cause the gain from making the mobile signals orthogonal to each other may exceed the loss in detection efficiency of the Di^K scheme. In order to satisfy the requirement for time alignment in orthogonal Walsh functions for the alternative mobile-to-cell link, each cell receiver determines the lime errtir froni nominal timing of each received signal. If a given received signal lags in timing, then the associated cell modulator and transmitter will transmit a command to this mobile to advance its transmit timing by a small increment. Conversely, if the received signal timing of a mobile leads the nominal liming, a command to retard by a small increment is Iraasmitled to the mobile. Tlie timing adjustment increments are made on the order of 1/8 PN chip or 101.7 nanoseconds. The commands are transmitted at a relatively low rate, on the order of 10 to 50 Hz and consist of a single bit inserted into the digital voice data flow. During a soft handoff operation, the mobile unit will-be receiving signals from two or more cells. Because the mobile unit can only align its timing in response to one of cells' timing adjust commands, the mobile unit will normally move its timing in response to the ctmimands received from iho slronj'.esl cell being received. The mobile I unit transmitted signal wiii thus Ix^ in time alignment with the cell with which it has the host p.ith. OthtTvyisc* )',reater mutual interference to other users will result. ; j If each cell receiver nwiving, h m(>l)ile sij'.nal performs the above j time error measurement and correction Ir.insmission operation, then all ) the mobiles' received si}*,nals Will normally be received with approximately the same timing, resulting in reduced interference. Figure 9 ilKislrales in block t iagram ft)rm an exemplary mobile imit CDMA telephone set. The mobile unit CDMA telephone set includes an antonnn 430 which is coupled through diplcxer 432 to analog receiver 344 and transmit power amplifier 436. Antenna 430 and diplexer 432 are of standard design and permit simultaneous transmission and reception through a single antenna. Antenna 430 collects transmitted signals and provides them through diplexer 432 to analog receiver 434. Receiver 434 receives the RF frequency signals from diplexer 432 which arc typically in the 850 MHz frequency band for amplification and frequency downconvcrsion to an IF frequency. This translation process is accomplished using a frequency synthesizer of standard design which permits the receiver to be tuned to any of the frequencies within the receive frequency band of the overall cellular telephone frequency banci.'ihe sijnials are also filtered and digitized for providing to digital da(a receivers 540 and 542 along with searcher receiver 544. The details of receiver 434 are further illustrated in Figure 10. Received signals from anl Tlie IF signals are onlpul from mixer 504 to bandpass filter (BPF) 508, typically a Surfaci- Acoiislic Wave (SAW) filter having a passband of approximately 1.25 MHz, vvh(>re Ihey arc from bandpa.ss filtered. The characteristics of the SAW filler are'cho.sen to match the waveform of the sigiuil transmitted by Ihe ,)I that is nunlulated by a FN sequence clocked at a fMi-determined rate^ which in the exemplary embodiment is 1.2288 Mlfz. This cjock rate is chosen to be an integer multiple of the baseband data rate of 4.6 kbps. The filtered signals are output from lilM- 508 as an input to a variable cain IF amplifier 510 where the signals arc again amplified. The amplified IF signals are output from IF amplifier 510 to analog to digita (A/D) converter 512 where the signals are digitized. The conversion o the IF signal to a digital signal occurs at a 9.8304 MMz clock rate in thi exemplary embodiment which is exactly eight times the FN chip rate Although (A/D) converter 512 is illustrated as part of receiver 534, i ct)uld instead be a part of the data and searcher receivers. Tlie digitizcc II- signals are are output from (A/D) converter 512 to data receivers 441 and 442, and .searcher receiver 444. Receiver 434 also performs a power control function for adjusting the transmit power of the mobile unit. An automatic gain control (AGC circuit 514 is also coupled to the output of IF amplifier 510. In respons( to the level of the amplified IF signal, AGC circuit 514 provides J feedback signal to the gain control input of IF amplifier 510. Receiver 43^ also uses AGC circuit 514 to generate an analog power control signal tha is provided to transmit power control circuitry 438. In Figure 9, the digitized signal output from receiver 434 i provided to digital data receivers 440 and 442 and to searcher receive 444. It should be understood that an inexpensive, low performano mobile unit might have only a single data receiver while highc performance units may have two or more to allow diversity reception. The digitized IF signal may contain the signals of many on-goin] calls together with the pilot carriers transmitted by the current cell-sit and all neighboring cell-sites. The function of the receivers 440 and 44 are to correlate the IF samples with the proper FN sequence. Thi correlation process provides a property that is well-known in the art a "processing gain" which enhances the signal-to-interference ratio of signal matching the proper FN sequence while not enhancing othe signals. Correlation output is theA synchronously detected using th pilot carrier from the closest cell-site as a carrier phase reference. Th result of this detection process is a sequence of encoded data symbols. A property of the FN sequencte as used in the present invention i that discrimination is provided agjainst multipath signals. When th signal arrives at the mobile receiver after passing through more tha one path, there will be a difference in the reception time of the signal. This reception lime difference corresponds to the difference in distance divided by the velocity of propagation. If this time difference exceeds one microsecond, then the correlation process will discriminate between the paths. The receiver can choose whether to track and receive the earlier or later path. If two receivers are provided, such as receivers 440 and 442, then two independent paths can be tracked and processed in parallel. Searcher receiver 444, under control of control processor 446 is for continuously scanning the time domain around the nominal time of a received pilot signal of the cell-site for other multi-path pilot signals from the same cell-site and for other cell-site transmitted pilot signals. Receiver 444 will measure the strength of any reception of a desired waveform at limes other than the nominal time. Receiver 444 compares signal strength in the received signals. Receiver 444 provides a signal strength signal to control processor 446 indicative of the strongest sij^nals. Processor 446 provides control signals to data receivers 440 and 442 for each to'process a different one of the strongest signals On occasion another cell-site transmitted pilot signal is of greater signal strength than the current cell-site signal strength. Control processor 446 then would generate a control message for transmission to the system controller via the current cell-site requesting a transfer of the cell to the cell-site corresponding to the strongest pilot signal. Receivers 440 and 442 may therefore handle calls through two different cell-sites. During a soft handoff operaliorl, the mobile unit will IH» receiving signals from two or more cells. Because the mobile unit can only align its liming in response to one of cells' timing adjust commands, the mobile unit wiil normally move its timing in response to the commands received from the strongest cell being received. The mobile unit transmitted.signal will thus be in time alignment wilh the cell with which it has the best path. Otherwise greater mutual interference to other users will result. Further details of an exemplary receiver, such as data receiver 440 is illustrated in further detail in Figure 10. Data receiver 440 includes PN generators 516 and 518 which generate the PNi and PNQ sequences in a manner and corresponding to those generated by the cell-site. Timing and sequence control signals are provided to PN generators 516 and 518 from control processor 446. Data receiver 440 also includes Walsh generator 520 which provides the appropriate Walsh function for communication with this mobile unit by the cell-site. Walsh generator 520 generates, in response to timing signals (not shown) and a function select signal from the control processor, a signal corresponding to an assigned Walsh sequence. The function select signal transmitted to the mobile unit by the cell-site as part of the call set up message. Tlie PN| and PNQ sequences output from PN generators 516 and 518 are respectively input to exclusive-OR gates 522 and 524. Walsh generator 520 provides its output to both of exclusive-OR gates 522 and 524 where the signals are exclusive-OR'ed and output the sequences PNj' and PNQ'. The sequences PNf and PNQ' are provided to receiver 440 where they are input to PN Ql^SK correlator 526. PN correlator 526 may be constructed in a manner similar to the PN correlator of the cell-site digital receivers. PN correlator 526 correlates the received 1 and Q cliannel data with the I*N|' and PNQ' secjuences and provides correlated I and Q channel data output to corresponding accumulators 528 and 530. Accumulators 528 and 530 accumulate the input information over a period of one symbol or 64 chips. The accumulator outputs are provided to pha.se rotator 532 which also receives a pilot pha.se signal from control processor 446. The phase of the received symbol data is rotated in .ucorciance wiUi liic phase of llu\ pilot signal as determined by the searcher receiver and the control -proce.ssor. The output from phase rotator 532 is the I channel data which is provided to the deinterleaver and decoder circuitry. Control processor 446 also i includes PN generator 534 which generates the user PN sequence ill response to an input mobile unit address or user ID. The PN sequence output from I'JNJ gener In the mobilo-to-cell link, user analog voice signals are typically, provided Ihrxjugh a handsel as an Mnput to baseband circuitry 450. Ba.scband circuitry 450 includes an analog lo digital (A/D) converter (not shown) which converts the analog signal to digital form. The digital signal is provided to the digital vocoder where it is encoded. The vocoder output is provided to a forward error correction (niC) encoding circuit (not shown) for error correction. In the exemplary embodiment the error correction encoding implemented is of a convolutional encoding scheme. The digitized encoded signal is output from baseband circuitry 450 to transmit modulator 452. Transmit modulator 452 first Walsh encodes the transmit data and then modulates the encoded signal on a PN carrier signal whose PN sequence is chosen according to the assigned address function for the call. The PN sequence is determined by control processor 446 from call setup information that is transmitted by the cell-site and decoded by receivers 440 and 442 and control processor 446. In the alternative, control processor 446 may determine the PN sequence through prearrangemcnt with the cell-site. Control processor 446 provides the PN sequence information to transmit modulator 452 and to receivers 440 and 442 for call dectKling. The output of transmit modulator 452 is provided to transmit power control circuitry 438. Signal transmission power is controlled by the Analog power control signal provided from receiver 434. Control bits transmitted by the cell-sites in the form power adjustment command are processed by data receivers 440 and 442. The power adjustment command is used by control processor 446 in setting the power level in mobile unit transtnission. In response to this command, control processor 446 generates a digital povyer control signal that is provided to circuitry 438. Further information bn the relationship of receivers 440 and 442, control processor 446 an^ transmit power control 438 with respect to power control is further described in the above-mentioned i copending patent application. Transmit power control circuitry 438 outputs the power controlled modulated signal to transmit power amplifier circuitry 436. Circuitry 436 amplifies and converts the IF signal io an RF frequency by mixing with a frequency synthesizer output signal jwhich tunes the signal to the proper output frequency. Circuitry 436 includes an amplifier which amplifies the power to a final output level. TT»e intended transmission signal is output from circuitry 436 to diplexer 432. Diplexer 432 couples the signal , to antenna 340 for transmission to the cell-sites. Control processor 446 also is capable of generating control messages such as cell-diversity mode requests and cell-site communication termination commands. These commands arc provided to transmit modulator 452 for transmission. Control processor 446 is responsive to the data received from data receivers 440 and 442, and search receiver 444 for making decisions relative to handoff and diversity combining. With respect to transmission by the mobile iniit, the mobile user analog voice signal is first passed through a digital vocoder. The vocoder output is then, in sequence, convolulional forward error correction (FEC) encoded, 64-ary orthogonal sequence encoded and modulated on a PN carrier signal. The 64-ary orthogonal sequence is generated by a Walsh function encoder. The encoder is controlled by collecting six successive binary symbol outputs from the convolutional FEC encoder. The six binary collectivtly determine which of the 64 possible Walsh sequences will be transmitted. TTie Walsh sequence is 64 bits long. Tlius, the Walsh "chip" rate must be 9600 » 3| * (1 /6) * 64 = 307200 Hz for a 9600 bps data transmission rale. | In the mobile-to-cell link, a common short PN sequence is used for all voice carriers in the system, while user address encoding is done using the user PN sequence generjator. The user PN secjuence is uniquely assigned to the mobile for an least the duration of the call. The user PN sequence is exclusive-OR'edJ with the common PN sequences, which are length 32768 augnu'nle|d maximal linear shift reg/ister sequences. Tlie resulting binary signials then each bi-phase modulate a quadrature carrier, are summed to forin a comp>osite signal, are bandpass filtered, and translated to an IF frecjuency output. In the exemplary embodiment, a p>ortion of the filteringiprocess is actually carried out by a i finite impulse response (FIR) digital filler operating on the binary sequence output. The modulator output is then power controlled by sig.nals from the digital control processor and the analog receiver, converted to the RF frequency of operation by mixing with a frequency synthesizer which tunes the signal to proper output frequency, and then amplified to the final output level. The transmit signal is then passed on to the diplexer and the antenna. Figure 11 illustrates a preferred, but yet exemplary, embodiment of mobile unit transmit modulator 452. Data is provided in digital form from the user digital baseband circuitry to encoder 600 where in the exemplary embodiment is convolutionally encoded. The output of encoder 600 is provided to interleaver 602 which in the exemplary embodiment is a block interleaver. The interleaved symbols are output from block interleaver 602 to Walsh encoder 604 of transmit modulator 452. Walsh encoder 604 utilizes the input symbols to generate a code sequence output. The Walsh sequence is provided to one input of exclusive-OR gate 606. Transmit modulator 452 Ivnthcr includes PN generator 608 which receives the mobile unit address as; an ijiput in determining the output PN sequence. PN generator 6()H; generates the user specific 42-bit sequence as was discussed with reference to Figure 3 and 4. A further attribute of PN generator 608 that is common to all user PN generators and not previously discussed is the use of a masking technique in generating the output user PN sequence. For example, a 42-bit mask is provided for that user with each bi|t of the 42-bit mask exclusive-OR'ed with a bit output from each register of the series of shift reg,ister that form the PN generator. The resuFs of the mask and shift register bit exclusive-OR operation are then eicclusive-OK'ed together to form the PN generator output that is used a^ the user PN sequence. The output PN sequence of PN generator 6P8, the sex^uence PNu, is input to exclusive-OR gale 606. The Walsh symbol data antl the PNy secjuence are exclusive-OR'ed in excIusivc-OR gate 606 and provided as in input to both of exclusive-OR gates 610 and 612. Transmit modulator 452 further includes PN generators 614 and 616 which respectively generate PNi and PNQ sequences. All mobile units use the same PNj and PNQ sequences. These PN sequences are in the exemplary embodiment the zero-shift used in the cell-to-mobile communications. The other input of cxclusivc-OR gates 610 and 612 are respectively provided with the PNi and I'NQ sequences output from PN generators 614 and 616. Tlie sequences PNj and PNQ are exclusive-OR'ed in the respective exclusive-OR gates with the output provided to transmit power control 438 (Figure 9). In the exemplary embodimcntj the mobilc-to-cell link uses rale r = 1/3 convolutional code with constraint length K = 9. The generators for the code are Ci = 557 (octal), Cj = 663 (octal), and C3 = 711 (octal). Similar to the cell-to-mobile link, code repetition is used to accommodate the four different data rates tliat the vocoder produces on a i 20 msec frame basis. Unlike the ccll-to-mobile link, the repealed code symbols are not transmitted over the:air at lower energy levels, rather only one code symbol of a repetition j^foup is transmilled nt the nominal power level. In conclusion, the colde "repetition in the exemplary embodiment is used merely as an expedient to fit the variable data rate scheme in the interleaving and moduljilion structure as it will be shown in the following paragraphs.- A block interleaver spanning 20 msec, exactly one vocoder frame, is used in the mobile-lo-cell link. Tl|\e numlx^r of code symlx^ls in 20 msec, a.ssuming a data rate of 9600 bps and a code rale r=l /3, is 576. The N and B parameters, N is equal to the nujmber of rows and B to the number of columns of the interleaver array .ire! 32 and IH, ri'.sp«.'Clivoly. (he code symbols are written into the interleavqr memory array by rows and read out by columns. | The modulation format is 64-ary orthogonal signalling. In other words, interleaved code symbols are grouped into groups of six to select one out of 64 orthogonal waveforms. The 64 time orthogonal waveforms are the same Walsh functions used as cover sequences in the cell-to-mobile link. The data modulation time interval is equal to 208.33 |.isec, and is referred to as a Walsh symbol interval. At 9600 bps, 208.33 |isec corresponds to 2 information bits and equivalenlly to 6 code symbols at a code symbol rate equal to 28800 sps. The Walsh symbol interval is subdivided into 64 equal length lime intervals, referred to as Walsh chips, each lasting 208.33 / 64 = 3.25 fuscc. The Walsh chip rale is then l/3.25p.sec = 307.2 kHz. Since the PN spreading rate is symmetric in the two links, i.e. 1.2288 MHz, there are exactly 4 PN chips per Walsh chip. A total of three PN generators are used in the mobile-to-cell link path. The user specific 42-bit PN generator and the pair of 15-bit I and Q channel PN generators. Following the user specific spreading operation^ the signal is QPSK spread as it was done in the ccll-to-mobilc link. Unlike the cell-to-mobile link, where each sector or celT was identified by unique sequences of length 2^^, hero all mobile units use the same I and Q PN sequences. Tliese PN sequences are the zero-shift sequences used in the cell-to-mobile b'nk, also referreid to as the pilot sequences. Code repetition and energy sc.iling arc used in the cell-to-mobile link to accommodate the variable rales produced by the vocoder. Tlie mobile-to-cell link uses a different scjieme based on a burst transmission. The vocoder produces four different data rale.s, i.e. 9600, 4800, 2400, and 1200 bps, on a 20 msec frame basis as in the cell-to-mobile link. The information bits are encoded by thci rale r = 1/3 convolulional encoder and code symbols are repeated 2, 4,jand 8 limes at the three lower data rates. Tlius, the code symbol rale is kbpt constant at 28800 sps. Pollowing the encoder, the code symbols are interleaved by the block inferleaver which spans exactly one viKoder fralme or 20 msec. A total of 576 code syrnbols am generated every 20 nisec by the Gonvolutional encoder; some of which might be repeated symbols. The code symbols sequence as it is Iransmilled is shown in Figure 12. Notice that a vocoder frame, 20 msec, has been subdivided into 16 i: slots each lasting 1.25 msec. The nuijnerology of the mobile-to-cell link is r i such that in each slot there are 36 code symbols al Ihe 28800 sps rale or equivalently 6 Walsh symbols at the 4800 sps rate. At the 1/2 rate, i.e. 4800 bps, the slots arc grouped into 8 groups each comprising 2 slots. At the 1/4 rate, i.e. 2400 bps, the slots are grouped into 4 groups each comprising 4 slots, and finally at the 1/8 rale, i.e. 1200 bps, tlic slots are grouped into 2 groups each comprising 8 slots. An exemplary symbol burst transmission pattern is further illustrated in Figure 12. For example, at the 1/4 rate, i.e. 2400 bps, during the fourth slot of the first group the fourth and eighth row of the interleaver memory array are read out by columns and sequentially transmitted. The slot position for the transmitted data must be randomiz<: in order to reduce the ihlorforonce.> The mobile-to-cell link timing is illustrated in Figure 13. Figure 13 expands upon the timing diagram of Figure 7 to include the mobilc- to-cell channels, i.e. voice and access. The synchronization of the mobile-to-cell link comprises the following steps: i 1. I3ea>dc successfully a syncifTicssago, i.e. CKC chock; 2. Load long PN shift register with state received in the sync message; and 3. Compensate for pilot code phase offset if receiving from a sector which uses a shifted pilot. j At this point the mobile ha^ complete synchronization, i.e. FN synchronization and real time synchronization, and can begin • to transmit on either the access chann The mobile unit in order to originate a call must be provided with signaling attributes in order to complete a call to another system user via a cell-site. In the mobile-to-cell link the envisioned access technique is the slotted ALOHA. An excmpl.iry transmission bit rale on llio reverse channel is 4800 bps. An access channel packet comprises of a preamble followed by the information. The preamble length is in liie exemplary embodiment an integer I-multiple of 20 msec frames and is a sector/cell parameter which the mobile receives in one of the pagihg channel messages. Since the cell receivers use the preambles to resolve propagation delays this scheme allows the preamble length to vary based on the coll radius. The users PN code for the access channel is either prearranged or transniitted to the mobile units on the paging channel. The modulation is fixed and constant for the duration of the preamble. The orthogonal waveform used in the preamble is Wo, i.e. the all zero Walsh function. Notice that all zero pallern at the input of the convolulional encoder generates the desired waveform W(). An access channel data packet may consist of one or at most two 20 msec frames. The coding, interleaving, and modulation of the access channel is exactly the same as for a voice channel at the 9600 bps rate. In an exemplary embodiment, the seclor/cell requires the mobile inuls to transmit a 40 msec preamble and the access channel message type requires one data frame. Let Np be the number of preamble frames where k is the number of 20 msec elapsed from a predefined lime origin. TTien mobiles are allowed to initiate transmission on the access channel only when the equation: (k, Np+2) = 0 is true. With respect to other communications applications it may be desirable to rearrange the various elements of the error correction coding, the orthogonal sequence coding and the PN coding to better fif the application. For example, in satellite mobile communications where the signals are relayed between large Hub earth stations and the mobile tern'tinals by one or more earth orbiting satellites, it may be desirable to employ coherent modulation and demodulation techniques in both directions of the link because the ch.umel is much more pha.se coherent than the terrestrial mobile channel. In such an application, the mobile modulator would not utilize m-.iry encotlinj; as described above. Instead, bi-phase or four-phase modulation of forward error correction syiTnbols might be employed with conventional coherent demodulation with carrier phase extracted from the received signal using Costns loop techniques. In addition, the orthogonal Walsh function channelization such as herein described for the cell-lo-mobile link may be employed. As long as me cnnnnel pnnsc remains reasonably coherent, this modulation and demodulation system provides operation with lower Eb/No than m-ary orthogonal signaling resulting in higher system capacity. In another embodiment, it may be preferable to encode the speech waveform directly into the RF waveform instead of utilizing a vocoder and FEC techniques. While the use of a vocoder and FEC techniques result in very high link performance, the complexity of implementation is high, resulting in additional cost and in high power consumption. These disadvantages may be especially unfavorable in a pocket portable telephone where battery consumption and cost are important. In customary digital telephone transmission practice, the speech waveform is represented in a digital format as 8 bit speech samples at a sample rate of 8 kHz. The CDMA system could encode the 8 bit samples directly into carrier phase angles. This would climinalo the need for n vocoder or a FEC encoder/decoder. It would also require a somewhat higher signal-to-noise ratio for good performance, resulting in lower capacity. In another alternative, the 8 bit speech samples could bo directly encoded into carrier amplitudes. In yet another alternative, the speech waveform samples could be encoded into carrier! phases and amplitudes. The previous description of the preferred embodiments is provided to enable any person skilled in the art to make or use the present invention. The various modification to these embotliinenls will be readily apparent to those sikilled in the art, and the generic principles defined herein may be applied to other embodiments without the use of the inventive faculty. Thus, the present invention is not intended to be limited to the embodiments shown herein but is to be accorded the widest scope consislent with the principles and novel features disclosed herein. ! WE-CLAIM: WE CLAIM : 1. A code division multiple access (CDMA) transmission system for spread spectrum modulation and transmission of a plurality of input digital user information signals each intended for a respective recipient user, said transmission system comprising spreading means for generating first and second spectrum spreading signals; pilot channel means for generating a pilot channel orthogonal function signal representative of a first orthogonal function selected from a set of orthogonal functions, combining said first and second spectrum spreading signals with said pilot channel orthogonal function signal, and providing as an output first and second pilot channel output signals; a plurality of user channel means each for receiving a respective one of a plurality of user information signals, generating a user cha'nnel orthogonal function signal representa ve of a selected one of said orthogonal functions of said set of orthogonal functions wherein each user channel means orthogonal function signal is of a different orthogonal function with respect to each other user channt orthogonal function signal and said pilot channel orthogonal function signals, combining said received user information signal with said generated user channel orthogonal function signal so as to provide a resultant user channel orthogonalised information signal, combining each resultant user channel orthogonalized information signal with said first and second spectrum spreading signals, and providing as an output from each respective user channel means corresponding first and second user channel output signals; transmission means for, receiving and converting said first and second pilot channel output signals to analog form, receiving and converting each user channel means first and second user channel output signals to analog form, combining said analog first pilot channel output signal and each analog first user channel output signal to provide a first combined signal, combining said analog second pilotichannel output signal and each analog second user channel output signal to provide a second combined signal, combining said first combined signal, with a first carrier signal so as to provide a first modulated carrier signal, combining said second combined signal with s second carrier signai so as to provide; a second modulated carriei signal,•combining said first and second modulated carrier signals as a composite modulated carrier signal and transmitting said composite modulated carrier signal', 2. The transmission system.; of claim 1 further comprising at least one ausxiliary channel means each for,receiving a respective auxiliary channel information signal, generating an auxiliary channel orthogonal functiion signal representative of a selected one of said orthogonal functions of said set of orthogonal functions wherein each auxiliary channel means orthogonal function signal is of a different orthogonal function ^ith respect to each other auxiliary channel means orthogonal function signal, each user channel orthogonal function signal and said pilot channel orthogonal function signal, combining said received auxiliary channel information signal with said generated auxiliary channel orthogonal function signal so as to provide a resultant auxiliary channel orthogonalized information signal, combining each auxiliary channel orthogonalized information signal, combining each auxiliary channel orthogonalized information signal with said first and second spectrum spreading signals, and providing as an output from each respective auxiliary channel means first and second auxiliary channel output signals to said transmission means; and said fcr transmission means further for, receiving and converting each auxiliary channel means first and second auxiliary channel output signals to analog foi..i, combining each analog first auxiliary channel output signal with said analog first pilot channel output signal and each analog first user channel output signal in said first combined signal, combining each analog second auxiliary channel output-signal with said analog second pilot channel output signal and each second user chamn^l output signal in said second combined signal. 3. The transmission system as claimed in claim 1 wherein said spreading means comprises first pseudorandom noise (PN) generator means for generating said first spectrum spreading signal of an In-Phase PN chip code; second PN generator means for generating said second spectrum spreading signal of a Quadrature-Phase PN chip code; and wherein said In-Phase and said Quadrature-Phase PN chip codes are each of a different polynominal function. 4. The transmission system of claim 3 wherein said pilot channel means comprises : pilot channel Walsh function generator mean" for generating said pilot ■ channel orthogonal function signal comprised of a Walsh functio;n chip sequence of zero state chips; pilot channel first combiiner means for receiver and i combining said first spectrum spreading signal with said pilot channel orthogonal function sig'najl, and providing said firtt pilot channel output signal; and pilot channel second combiner means for receiving & .' combining ^aid second spectrum spreading signal • with said pilot channel orthogonal function signal, and providing said second pilot channei. output signal. .. . . „. i I 5. The transmission system as claimed in claim 4 whereir each user channel means comprise!? user channel Walsh functior generator means for generating said respective user channel orthogonal function signal comprised of a selected Walsh function chip sequence of zero and one state chips; user channel first combiner means for receiving and combining said respective user information signal with said generated user channel orthogonal function signal, and providing said user channel orthogonalised information signal; user channel second combiner means for receiving and combining said first spectrum spreading signal with said user channel orthogonalised information signal, and providing said first user channel output signal; and user channel third combiner means for receiving and combining said second spectrum spreading signal with said generated user ohanne orthogonalized information signal, and providing said second user channel output signal. f G. The transmission system as claimed in claim 5, wherein each auxiliary channel means comprises auxiliary chanriel Walsh i function generator means for generating said respective auxiliary channel orthogonal function signal |comprised of a selected Walsh function chip sequence of zero arjd one state chips; auxiliary channel first combiner means for [receiving and combining said respective auxiliary _ lpfo^^'*ation i signal with said generated auxiliary channel orthogonal function signal, and providing said i ■* auxiliary channel orthogonalized information signal; auxiliary channel second combiner means for receiving and combining said first spectrum spreading signal with said auxiliary channel orthogonalized information signal, and providing said first auxiliary channel output signal, and auxiliary channel third combiner means for receiving and combining said second spectrum spreading signal with said generated auxiliary channel orthogonalized information signal, and providing said second auxiliary channel output signal. 7. A code division multiple access (CDMA) transmission system substantially as hereinbefore described with reference to the accompanying dra.wings. |
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1541-mas-1995 correspondence-others.pdf
1541-mas-1995 correspondence-po.pdf
1541-mas-1995 description (complete).pdf
| Patent Number | 189670 | |||||||||||||||||||||
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| Indian Patent Application Number | 1541/MAS/1995 | |||||||||||||||||||||
| PG Journal Number | 30/2009 | |||||||||||||||||||||
| Publication Date | 24-Jul-2009 | |||||||||||||||||||||
| Grant Date | 27-Feb-2004 | |||||||||||||||||||||
| Date of Filing | 27-Nov-1995 | |||||||||||||||||||||
| Name of Patentee | M/S. QUALCOMM INC | |||||||||||||||||||||
| Applicant Address | 10555 SORRENTO VALLEY ROAD, SAN DIEGO, CA 92121, | |||||||||||||||||||||
Inventors:
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| PCT International Classification Number | H04L 27/18 | |||||||||||||||||||||
| PCT International Application Number | N/A | |||||||||||||||||||||
| PCT International Filing date | ||||||||||||||||||||||
PCT Conventions:
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